Power loop control based envelope tracking

ABSTRACT

Configuration-feedback circuitry and transceiver circuitry are disclosed. The configuration-feedback circuitry regulates an output power from a radio frequency power amplifier based on a difference between a target output power from the radio frequency power amplifier and a measured output power from the radio frequency power amplifier. The transceiver circuitry regulates a modulated power supply voltage, which is used by the radio frequency power amplifier to provide power for amplification, based on the difference between the target output power from the radio frequency power amplifier and the measured output power from the radio frequency power amplifier.

RELATED APPLICATIONS

The present application claims priority to and is a continuation-in-partof International Patent Application No. PCT/US12/36858, filed May 7,2012, entitled “POWER MANAGEMENT SYSTEM FOR PSEUDO-ENVELOPE AND AVERAGEPOWER TRACKING,” which claims priority to U.S. Provisional PatentApplications No. 61/482,702, filed May 5, 2011; No. 61/484,613, filedMay 10, 2011; No. 61/508,202, filed Jul. 15, 2011; No. 61/530,625, filedSep. 2, 2011; No. 61/544,051, filed Oct. 6, 2011; No. 61/551,605, filedOct. 26, 2011; No. 61/565,138, filed Nov. 30, 2011; and No. 61/576,520,filed Dec. 16, 2011.

International Patent Application No. PCT/US12/36858, filed May 7, 2012,claims priority to and is a continuation-in-part of U.S. patentapplication Ser. No. 13/218,400, filed Aug. 25, 2011, entitled “BOOSTCHARGE-PUMP WITH FRACTIONAL RATIO AND OFFSET LOOP FOR SUPPLYMODULATION,” now U.S. Pat. No. 8,519,788, which was also filed asInternational Patent Application No. PCT/US11/49243 on Aug. 25, 2011.

International Patent Application No. PCT/US12/36858, filed May 7, 2012,claims priority to and is a continuation-in-part of International PatentApplication No. PCT/US11/54106, filed Sep. 29, 2011, entitled “SINGLEμC-BUCKBOOST CONVERTER WITH MULTIPLE REGULATED SUPPLY OUTPUTS.”

International Patent Application No. PCT/US12/36858, filed May 7, 2012,claims priority to and is a continuation-in-part of U.S. patentapplication Ser. No. 13/316,229, filed Dec. 9, 2011, entitled“PSEUDO-ENVELOPE FOLLOWER POWER MANAGEMENT SYSTEM WITH HIGH FREQUENCYRIPPLE CURRENT,” now U.S. Pat. No. 8,633,766, which was also filed asInternational Patent Application No. PCT/US11/64255 on Dec. 9, 2011.U.S. patent application Ser. No. 13/316,229, filed Dec. 9, 2011, is acontinuation-in-part of U.S. patent application Ser. No. 13/218,400,filed Aug. 25, 2011, which was also filed as International PatentApplication No. PCT/US11/49243 on Aug. 25, 2011.

International Patent Application No. PCT/US12/36858, filed May 7, 2012,claims priority to and is a continuation-in-part of U.S. patentapplication Ser. No. 13/367,973, filed Feb. 7, 2012, entitled “GROUPDELAY CALIBRATION METHOD FOR POWER AMPLIFIER ENVELOPE TRACKING,” nowU.S. Pat. No. 8,942,313, which was also filed as International PatentApplication No. PCT/US12/24124 on Feb. 7, 2012.

International Patent Application No. PCT/US12/36858, filed May 7, 2012,claims priority to and is a continuation-in-part of U.S. patentapplication Ser. No. 13/423,649, filed Mar. 19, 2012, entitled“APPARATUSES AND METHODS FOR RATE CONVERSION AND FRACTIONAL DELAYCALCULATION USING A COEFFICIENT LOOK UP TABLE, ” now U.S. Pat. No.8,624,760.

International Patent Application No. PCT/US12/36858, filed May 7, 2012,claims priority to and is a continuation-in-part of U.S. patentapplication Ser. No. 13/363,888, filed Feb. 1, 2012, entitled “FASTENVELOPE SYSTEM CALIBRATION,” now U.S. Pat. No. 8,611,402, which wasalso filed as International Patent Application No. PCT/US12/23495, onFeb. 1, 2012.

All of the applications listed above are hereby incorporated herein byreference in their entireties.

FIELD OF THE DISCLOSURE

The embodiments described herein relate to a power management system fordelivering current to a linear radio frequency power amplifier. Moreparticularly, the embodiments relate to the use of a pseudo-envelopetracker in a power management system of mobile communications equipment.

BACKGROUND

Next-generation mobile devices are morphing from voice-centrictelephones to message and multimedia-based “smart” phones that offerattractive new features. As an example, smart phones offer robustmultimedia features such as web-browsing, audio and video playback andstreaming, email access and a rich gaming environment. But even asmanufacturers race to deliver ever more feature rich mobile devices, thechallenge of powering them looms large.

In particular, the impressive growth of high bandwidth applications forradio frequency (RF) hand-held devices has led to increased demand forefficient power saving techniques to increase battery life. Because theradio frequency power amplifier of the mobile device consumes a largepercentage of the overall power budget of the mobile device, variouspower management systems have been proposed to increase the overallpower efficiency of the radio frequency power amplifier. As such, thereremains a need to further improve the power efficiency of mobile devicesto provide extended battery life. In this regard, there is a need toimprove the power management system of mobile devices.

SUMMARY

Configuration-feedback circuitry and transceiver circuitry aredisclosed, according to one embodiment of the present disclosure. Theconfiguration-feedback circuitry regulates an output power from a radiofrequency power amplifier based on a difference between a target outputpower from the radio frequency power amplifier and a measured outputpower from the radio frequency power amplifier. The transceivercircuitry regulates a modulated power supply voltage, which is used bythe radio frequency power amplifier to provide power for amplification,based on the difference between the target output power from the radiofrequency power amplifier and the measured output power from the radiofrequency power amplifier.

By regulating the modulated power supply voltage based on the differencebetween the target output power from the radio frequency power amplifierand the measured output power from the radio frequency power amplifier,degradation of an output signal from the radio frequency power amplifiermay be reduced or avoided.

Those skilled in the art will appreciate the scope of the disclosure andrealize additional aspects thereof after reading the following detaileddescription in association with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings incorporated in and forming a part of thisspecification illustrate several aspects of the disclosure, and togetherwith the description serve to explain the principles of the disclosure.

FIG. 1A depicts an embodiment of a pseudo-envelope follower powermanagement system for managing power supplied to a linear radiofrequency power amplifier according to one embodiment of thepseudo-envelope follower power management system.

FIG. 1B depicts an embodiment of the pseudo-envelope follower powermanagement system for managing power supplied to a linear radiofrequency power amplifier according to an alternate embodiment of thepseudo-envelope follower power management system.

FIG. 2A depicts one embodiment of a pseudo-envelope tracking modulatedpower supply system according to one embodiment of the pseudo-envelopetracking modulated power supply system.

FIG. 2B depicts one embodiment of a pseudo-envelope tracking modulatedpower supply system according to an alternate embodiment of thepseudo-envelope tracking modulated power supply system.

FIG. 2C depicts one embodiment of a pseudo-envelope tracking modulatedpower supply system according to an additional embodiment of thepseudo-envelope tracking modulated power supply system.

FIG. 3 depicts a method for operating the pseudo-envelope trackingmodulated power supply system according to one embodiment of thepseudo-envelope tracking modulated power supply system.

FIG. 4 depicts one embodiment of a set of iso-gain contours of a radiofrequency power amplifier.

FIG. 5 depicts one embodiment of a portion of a communication deviceincluding various embodiments of an envelope tracking modulated powersupply system with transceiver power control loop compensation.

FIG. 6 depicts one embodiment of transceiver circuitry.

FIG. 7A depicts one embodiment of a saw tooth effect.

FIG. 7B depicts an alternate embodiment of the saw tooth effect.

FIG. 7C depicts a method for reducing the saw tooth effect.

FIGS. 8A-D depict a method for compensating a control signal generatedas part of an envelope tracking system.

DETAILED DESCRIPTION

The embodiments set forth below represent the necessary information toenable those skilled in the art to practice the disclosure andillustrate the best mode of practicing the disclosure. Upon reading thefollowing description in light of the accompanying drawings, thoseskilled in the art will understand the concepts of the disclosure andwill recognize applications of these concepts not particularly addressedherein. It should be understood that these concepts and applicationsfall within the scope of the disclosure and the accompanying claims.

Configuration-feedback circuitry and transceiver circuitry aredisclosed, according to one embodiment of the present disclosure. Theconfiguration-feedback circuitry regulates an output power from a radiofrequency power amplifier based on a difference between a target outputpower from the radio frequency power amplifier and a measured outputpower from the radio frequency power amplifier. The transceivercircuitry regulates a modulated power supply voltage, which is used bythe radio frequency power amplifier to provide power for amplification,based on the difference between the target output power from the radiofrequency power amplifier and the measured output power from the radiofrequency power amplifier.

By regulating the modulated power supply voltage based on the differencebetween the target output power from the radio frequency power amplifierand the measured output power from the radio frequency power amplifier,degradation of an output signal from the radio frequency power amplifiermay be reduced or avoided.

FIG. 1A depicts an embodiment of a pseudo-envelope follower powermanagement system 10A for managing power supplied to a linear radiofrequency power amplifier 22 according to one embodiment of thepseudo-envelope follower power management system 10A. FIG. 1B depicts anembodiment of a pseudo-envelope follower power management system 10B formanaging power supplied to the linear radio frequency power amplifier 22according to an alternate embodiment of the pseudo-envelope followerpower management system 10A. As such, FIGS. 1A and 2A depict an exampleembodiment of the pseudo-envelope follower power management system 10Aincluding a multi-level charge pump buck converter 12, a parallelamplifier circuit 14, a power inductor 16, a coupling circuit 18, and abypass capacitor 19. The bypass capacitor 19 has a bypass capacitorcapacitance, C_(BYPASS).

The multi-level charge pump buck converter 12 and the parallel amplifiercircuit 14 may be configured to operate in tandem to generate a poweramplifier supply voltage, V_(CC), at a power amplifier supply output 28of the pseudo-envelope follower power management system 10A for thelinear radio frequency power amplifier 22. The power amplifier supplyvoltage, V_(CC), may also be referred to as a modulated power supplyvoltage, V_(CC). The power amplifier supply output 28 provides an outputcurrent, I_(OUT), to the linear radio frequency power amplifier 22. Thelinear radio frequency power amplifier 22 may include a radio frequencypower amplifier input configured to receive a modulated radio frequencyinput signal having an input power P_(IN). The linear radio frequencypower amplifier 22 may further include a radio frequency power amplifieroutput coupled to an output load, Z_(LOAD). The linear radio frequencypower amplifier 22 may generate an amplified modulated radio frequencyoutput signal having an output power P_(OUT) in response to themodulated radio frequency input signal having the input power P_(IN).

As an example, the output load, Z_(LOAD), may be an antenna. The radiofrequency power amplifier output may generate the amplified modulatedradio frequency output signal as a function of the modulated radiofrequency input signal and the power amplifier supply voltage, V_(CC).In some embodiments, the power amplifier supply voltage, V_(CC), may bemodulated to substantially follow the signal envelope characteristic ofthe modulated radio frequency input signal to improve the powerefficiency of the pseudo-envelope follower power management system 10A.The amplified modulated radio frequency output signal may be provided tothe antenna for transmission. The multi-level charge pump buck converter12 may include a supply input 24, (V_(BAT)), configured to receive abattery voltage, V_(BAT), from a battery 20 and a switching voltageoutput 26 configured to provide a switching voltage, V_(SW). Theswitching voltage output 26 may be coupled to the power amplifier supplyoutput 28 by the power inductor 16, where the power inductor 16 couplesto the bypass capacitor 19 to form an output filter 29 for the switchingvoltage output 26 of the multi-level charge pump buck converter 12. Assuch, the power inductor 16 is coupled between the switching voltageoutput 26 and the power amplifier supply output 28. The power inductor16 provides a power inductor current, I_(SW) _(—) _(OUT), to the poweramplifier supply output 28. The parallel amplifier circuit 14 mayinclude a parallel amplifier supply input 30 configured to receive thebattery voltage, V_(BAT), from the battery 20, a parallel amplifieroutput 32A, a first control input 34 configured to receive a V_(RAMP)signal, and a second control input configured to receive the poweramplifier supply voltage, V_(CC). The parallel amplifier output 32A ofthe parallel amplifier circuit 14 may be coupled to the power amplifiersupply voltage V_(CC), by the coupling circuit 18. A parallel amplifieroutput voltage, V_(PARA) _(—) _(AMP), is provided by the parallelamplifier circuit 14.

As an example, the parallel amplifier circuit 14 may generate theparallel amplifier output voltage, V_(PARA) _(—) _(AMP), based on thedifference between the V_(RAMP) signal and the power amplifier supplyvoltage, V_(CC). Thus, the V_(RAMP) signal may represent either ananalog or digital signal that contains the required supply modulationinformation for a power amplifier collector of a linear radio frequencypower amplifier. Typically, the V_(RAMP) signal is provided to theparallel amplifier circuit 14 as a differential analog signal to providecommon mode rejection against any noise or spurs that could appear onthis signal. The V_(RAMP) signal may be a time domain signal, V_(RAMP)(t), generated by a transceiver or modem and used to transmit radiofrequency signals. For example, the V_(RAMP) signal may be generated bya digital baseband processing portion of the transceiver or modem, wherethe digital V_(RAMP) signal, V_(RAMP) _(—) _(DIGITAL), isdigital-to-analog converted to form the V_(RAMP) signal in the analogdomain. In some embodiments, the “analog” V_(RAMP) signal is adifferential signal. The transceiver or a modem may generate theV_(RAMP) signal based upon a known radio frequency modulation Amp(t)*cos(2*pi*f_(RF)*t+Phase (t)). The V_(RAMP) signal may represent thetarget voltage for the power amplifier supply voltage, V_(CC), to begenerated at the power amplifier supply output 28 of the pseudo-envelopefollower power management system 10A, where the pseudo-envelope followerpower management system 10A provides the power amplifier supply voltage,V_(CC), to the linear radio frequency power amplifier 22. Also theV_(RAMP) signal may be generated from a detector coupled to the linearradio frequency power amplifier 22.

For example, the parallel amplifier circuit 14 includes the parallelamplifier output 32A that provides the parallel amplifier outputvoltage, V_(PARA) _(—) _(AMP), to the coupling circuit 18. The parallelamplifier output 32A sources a parallel amplifier circuit outputcurrent, I_(PAWA) _(—) _(OUT), to the coupling circuit 18. The parallelamplifier circuit 14, depicted in FIG. 1A and FIG. 1B, may provide aparallel amplifier circuit output current estimate 40, I_(PAWA) _(—)_(OUT) _(—) _(EST), to the multi-level charge pump buck converter 12 asan estimate of the parallel amplifier circuit output current I_(PAWA)_(—) _(OUT), of the parallel amplifier circuit 14. Thus, the parallelamplifier circuit output current estimate 40, I_(PAWA) _(—) _(OUT) _(—)_(EST), represents an estimate of the parallel amplifier circuit outputcurrent I_(PAWA) _(—) _(OUT), provided by the parallel amplifier circuit14 as a feedback signal to the multi-level charge pump buck converter12. Based on the parallel amplifier circuit output current estimate 40,I_(PAWA) _(—) _(OUT) _(—) _(EST), the multi-level charge pump buckconverter 12 may be configured to control the switching voltage, V_(SW),provided at the switching voltage output 26 of the multi-level chargepump buck converter 12.

In some embodiments of the pseudo-envelope follower power managementsystem 10A, depicted in FIG. 1A, and the pseudo-envelope follower powermanagement system 10B, depicted in FIG. 1B, the coupling circuit 18 maybe an offset capacitor, C_(OFFSET). An offset voltage, V_(OFFSET), maybe developed across the coupling circuit 18. In other alternativeembodiments, the coupling circuit 18 may be a wire trace such that theoffset voltage, V_(OFFSET), between the parallel amplifier outputvoltage, V_(PARA) _(—) _(AMP), and the power amplifier supply voltageoutput, V_(CC), is zero volts. In still other embodiments, the couplingcircuit may be a transformer.

FIG. 2A depicts one embodiment of a pseudo-envelope tracking modulatedpower supply system 868 according to one embodiment of thepseudo-envelope tracking modulated power supply system 868.

FIG. 2B depicts one embodiment of a pseudo-envelope tracking modulatedpower supply system 868 according to an alternate embodiment of thepseudo-envelope tracking modulated power supply system 868.

FIG. 2C depicts one embodiment of a pseudo-envelope tracking modulatedpower supply system 868 according to an additional embodiment of thepseudo-envelope tracking modulated power supply system 868.

FIG. 2A depicts a high level illustration of a pseudo-envelope trackingmodulated power supply system 868 that may include a radio frequencypower amplifier 869 configured to be powered by or under the control ofa power management system 870. The power amplifier supply voltage,V_(CC), may also be referred to as a modulated power supply voltage,V_(CC), that is generated at a modulated power supply output 876.

Similar to other previously described switch mode power supplyconverters, multi-level charge pump buck converters, and parallelamplifier circuits, a switch mode power supply converter 872 and aparallel amplifier circuit 874 may be configured to receive an inputsupply voltage from a battery 20. The battery 20 may provide a supplyvoltage substantially equal to the battery voltage, V_(BAT).

For the sake of simplicity of description, and not by way of limitation,the pseudo-envelope tracking modulated power supply system 868 mayinclude a controller 50 coupled via a control bus 44 to the switch modepower supply converter 872 and the parallel amplifier circuit 874. Theswitch mode power supply converter 872 may be arranged to cooperativelyoperate with the parallel amplifier circuit 874 to form the powermanagement system 870, which generates the modulated power supplyvoltage, V_(CC), at the modulated power supply output 876. For example,in some embodiments, the controller 50 may configure the switch modepower supply converter 872 and the parallel amplifier circuit 874 tooperate in various power level modulation modes, depending upon theexpected output power to be generated by the radio frequency poweramplifier 869 during a transmission of information. In some embodiments,the control functions described with respect to the controller 50 may beincorporated into a digital baseband modem or transceiver circuit thatprovides a differential V_(RAMP) signal as a control signal to the powermanagement system 870 based on a radio frequency input signal providedto the radio frequency power amplifier 869 for transmission.

The power management system 870 may be configured to receive adifferential V_(RAMP) signal having a non-inverted V_(RAMP) signalcomponent, V_(RAMP)+, and an inverted V_(RAMP) signal component,V_(RAMP)−. In some alternative embodiments, the power management system870 may be configured to receive a V_(RAMP) signal that is a singleended V_(RAMP) signal, a differential V_(RAMP) signal, and/or both thesingle ended V_(RAMP) signal and the differential V_(RAMP) signal. Thedifferential V_(RAMP) signal may be provided as a control signal togovern generation of the modulated power supply voltage, V_(CC).Illustratively, the switch mode power supply converter 872 and theparallel amplifier circuit 874 may each be configured to receive thedifferential V_(RAMP) signal.

The switch mode power supply converter 872 may include the switchingvoltage output 26 in communication with the modulated power supplyoutput 876. A power inductor 16 may be coupled between the switchingvoltage output 26 and the modulated power supply output 876.

The parallel amplifier circuit 874 may include a parallel amplifiercircuit output 874A in communication with the modulated power supplyoutput 876. Illustratively, in some embodiments, a coupling capacitor18A is coupled between the modulated power supply output 876 and theparallel amplifier circuit output 874A. In addition, the parallelamplifier circuit 874 may also include a first linear regulator output874B, LDO₁ OUTPUT, in communication with the modulated power supplyoutput 876. In some power level modulation modes, the power managementsystem 870 may configure the first linear regulator output 874B, LDO₁OUTPUT, to provide a high impedance path with respect to ground. Inother power level modulation modes, the power management system 870 mayconfigure the first linear regulator output 874B, LDO₁ OUTPUT, to applya first linear regulator output current 878A, I_(LDO), to the modulatedpower supply output 876 to regulate the modulated power supply voltage,V_(CC).

The parallel amplifier circuit 874 may also include a second linearregulator output 874D, LDO₂ OUTPUT, in communication with a CMOS logicsupply input 869C. In some power level modulation modes, the powermanagement system 870 may configure the second linear regulator output874D, LDO₂ OUTPUT, to provide a second linear regulator output voltage,V_(LDO2), to the CMOS logic supply input 869C as a function of thebattery voltage, V_(BAT). As an example, the CMOS logic supply input869C may include a minimum CMOS logic supply voltage threshold.Accordingly, the power management system 870 may configure the secondlinear regulator output 874D, LDO₂ OUTPUT, to provide a second linearregulator output voltage, V_(LDO2), to the CMOS logic supply input 869Cthat is at least equal to the minimum CMOS logic supply voltagethreshold.

The parallel amplifier circuit 874 may also include a switch mode powersupply converter control output 874E configured to output a switch modepower supply feedback signal 40A to the switch mode power supplyconverter 872, depicted in FIG. 2B. In addition, the parallel amplifiercircuit 874 may provide a threshold offset current 42, I_(THRESHOLD)_(—) _(OFFSET), to the switch mode power supply converter 872. Theparallel amplifier circuit 874 may receive the modulated power supplyvoltage, V_(CC), from the modulated power supply output 876 at amodulated power supply voltage feedback input 874F. The parallelamplifier circuit 874 may use the input to the modulated power supplyvoltage feedback input 874F as a feedback signal to regulate themodulated power supply voltage, V_(CC).

In some embodiments, the radio frequency power amplifier 869 may be alinear radio frequency power amplifier. The radio frequency poweramplifier 869 may include a radio frequency power amplifier inputconfigured to receive a modulated radio frequency input signal from adigital baseband processing portion of the transceiver or modem, wherethe modulated radio frequency input signal has an input power, P_(IN).In addition, the radio frequency power amplifier 869 may also include aradio frequency power amplifier output in communication with an antennavia the radio frequency duplexer and switch (not shown). The radiofrequency power amplifier 869 may generate an amplified modulated radiofrequency output signal having an output power P_(OUT) at the radiofrequency power amplifier output.

In some embodiments, the radio frequency power amplifier 869 may includea collector voltage supply input 869A configured to receive themodulated power supply voltage, V_(CC), from the modulated power supplyoutput 876. The radio frequency power amplifier 869 may further includea battery voltage supply input 869B configured to receive the batteryvoltage, V_(BAT), from the battery 20. The radio frequency poweramplifier 869 may also include a CMOS logic supply input 869C configuredto receive a second linear regulator output voltage, V_(LDO2).

In some embodiments of the power management system 870, the powermanagement system 870 may be configured to operate in various powerlevel modulation modes based on an expected output power to be generatedby the radio frequency power amplifier 869 during a data transmission.In addition, the power management system 870 may be reconfigured priorto each data transmission to minimize the energy drawn from the battery20 during the data transmission. For example, some embodiments of thepower management system 870 may be configured to operate in one of manypower level modulation modes on a data transmission slot by datatransmission slot basis.

As a non-limiting list of example power level modulation modes ofoperation, the power management system 870 may be configured to operatein a set of power level modulation modes including a high powermodulation mode, a medium power modulation mode, and low powermodulation mode. As another example, in other embodiments, the powermanagement system 870 may be configured to operate in a set of powerlevel modulation modes including a high power modulation mode, a mediumpower modulation mode, a medium power average power tracking modulationmodes, and a low power average power tracking modulation mode. In otherembodiments, the medium power average power tracking modulation modesare omitted.

As a further example, in a slow tracking mode of operation, the powermanagement system 870 may disable the switch mode power supply converter872, and configure the parallel amplifier circuit 874 to track anenvelope of a modulated radio frequency input signal to be transmittedby the radio frequency power amplifier 869 as a function of a slowlymodulated or unmodulated differential V_(RAMP) signal. In the notracking mode of operation, the power management system 870 may furtherconfigure the parallel amplifier circuit 874 to provide a modulatedpower supply voltage, V_(CC), based on a substantially unmodulateddifferential V_(RAMP) signal. In the no tracking mode, the powermanagement system 870 may be configured to generate a modulated powersupply voltage, V_(CC) that has a substantially fixed voltage level. Inthe slow tracking mode of operation, the parallel amplifier circuit 874may be configured to slowly track the envelope of the differentialV_(RAMP) signal.

To minimize energy consumed from the battery during a data transmission,the power management system 870 may enable and disable various elementsand signal processing functions of the switch mode power supplyconverter 872 and the parallel amplifier circuit 874 as a function ofthe power level modulation mode of operation. In some embodiments, thepower management system 870 may disable the least energy efficientelement and signal processing functions as a function of the expectedoutput power to be generated by the radio frequency power amplifier 869.For example, the power management system 870 may disable portions of theswitch mode power supply converter 872, the parallel amplifier circuit874, and/or a combination thereof as the expected output power of theradio frequency power amplifier 869 decreases to achieve an overalldecrease in energy consumption from the battery 20 during a datatransmission. In addition, some embodiments of the power managementsystem 870 may generate the modulated power supply output 876 using aless energy efficient device or power generation circuit in response toan expected output power of the radio frequency power amplifier 869falling below a low power modulation mode threshold in order to disableother energy consuming circuitry and achieve an overall reduction inenergy drawn from the battery 20 during a data transmission.

As a non-limiting example, in some embodiments of the high power levelmodulation mode and the medium power modulation mode, the powermanagement system 870 may configure the switch mode power supplyconverter 872 and the parallel amplifier circuit 874 to generate themodulated power supply voltage, V_(CC), based on the differentialV_(RAMP) signal as a function of the expected output power to begenerated by the radio frequency power amplifier 869 during the datatransmission. As an example, in some embodiments of the high powermodulation mode and the medium power modulation mode, the parallelamplifier circuit 874 may be configured to govern the operation of theswitch mode power supply converter 872, regulate generation of themodulated power supply voltage, V_(CC), as a function of thedifferential V_(RAMP) signal, and regulate the offset voltage,V_(OFFSET), across the coupling capacitor 18A. In one embodiment of theswitch mode power supply converter 872, during the high power modulationmode, the switch mode power supply converter 872 boosts the poweramplifier supply voltage, V_(CC), above the battery voltage, V_(BAT).

In general, in one embodiment of the pseudo-envelope tracking modulatedpower supply system 868, during the high power modulation mode and themedium power modulation mode, the power amplifier supply voltage,V_(CC), is modulated and provides envelope tracking. Further, during thelow power average power tracking mode, the power amplifier supplyvoltage, V_(CC), is not modulated and provides average power tracking.In this regard, during the low power average power tracking mode, amagnitude of the power amplifier supply voltage, V_(CC), is adjustablebased on an expected output power from the radio frequency poweramplifier 869.

As an example, the parallel amplifier circuit 874 may be configured toapply a parallel amplifier circuit output current, I_(PAWA) _(—) _(OUT),from the parallel amplifier circuit 874 to regulate the modulated powersupply voltage, V_(CC), as a function of the differential V_(RAMP)signal. The parallel amplifier circuit 874 may provide the switch modepower supply feedback signal 40A and a threshold offset current 42,I_(THRESHOLD) _(—) _(OFFSET), to govern the operation of the switch modepower supply converter 872 and regulate the offset voltage, V_(OFFSET),across the coupling capacitor 18A. In response to the switch mode powersupply feedback signal 40A and the threshold offset current 42,I_(THRESHOLD) _(—) _(OFFSET), and the differential V_(RAMP) signal, theswitch mode power supply converter 872 may generate the switchingvoltage, V_(SW), at the switching voltage output 26, and provide adelayed I_(COR) estimated switching voltage output 38C, V_(SW) _(—)_(EST) _(—) _(DELAY) _(—) _(ICOR), to the parallel amplifier circuit874. In addition, the parallel amplifier circuit 874 may configure thefirst linear regulator output 874B, LDO₁ OUTPUT, to provide a highimpedance path to ground. Depending on the battery voltage, V_(BAT), theparallel amplifier circuit 874 may configure the second linear regulatoroutput 874D, LDO₂ OUTPUT, to provide the second linear regulator outputvoltage, V_(LDO2), at least equal to the minimum CMOS logic supplyvoltage to the CMOS logic supply input 869C of the radio frequency poweramplifier 869.

As another non-limiting example, in some embodiments of the low powermodulation mode, the power management system 870 may disable the switchmode power supply converter 872 and configure the switching voltageoutput 26 to provide a high impedance path to ground. In addition, theparallel amplifier circuit output 874A may be configured to provide alow impedance path to ground to short the negative terminal of thecoupling capacitor 18A to ground.

To generate the modulated power supply voltage, V_(CC), the first linearregulator output 874B, LDO₁ OUTPUT, may be configured to apply a linearregulator output current, I_(LDO), to the modulated power supply output876 as a function of the differential V_(RAMP) signal and a selectedtracking mode of operation. The differential V_(RAMP) signal may besubstantially constant or only slowly modulated during the datatransmission. In the case where the differential V_(RAMP) signal issubstantially constant, the power management system 870 may configurethe parallel amplifier circuit 874 to operate in a no tracking mode ofoperation. In the no tracking mode of operation, the first linearregulator output 874B, LDO₁ OUTPUT, may regulate the modulated powersupply voltage, V_(CC), to be substantially constant or unmodulatedduring data transmission. Alternatively, in the case where thedifferential V_(RAMP) signal slowly changes relative to the envelope ofthe radio frequency input signal provided to the radio frequency poweramplifier 869, the power management system 870 may configure the firstlinear regulator output 874B, LDO₁ OUTPUT, to slowly track thedifferential V_(RAMP) signal during the data transmission.

In some embodiments of the low power modulation mode, the parallelamplifier circuit 874 may disable the second linear regulator output874D, LDO₂ OUTPUT. In some alternative embodiments of the low powermodulation mode, the parallel amplifier circuit 874 may configure thesecond linear regulator output 874D, LDO₂ OUTPUT, to selectively outputthe second linear regulator output voltage, V_(LDO2), to the CMOS logicsupply input 869C as a function of the battery voltage, V_(BAT).

FIG. 2B depicts a non-limiting example of the embodiments of the switchmode power supply converter 872 that may be selectively configured tooperate in a number of various buck converter modes, a number of variousenvelope tracking modes, a number of various average power trackingmodes, and/or a combination thereof as a function of an expected outputpower to be generated by the radio frequency power amplifier 869depicted in FIGS. 2A and 2C. The switch mode power supply converter 872may further include an off mode. The various embodiments of the switchmode power supply converter 872 are described with continuing referenceto the various embodiments of the power management system 870, depictedin FIGS. 2A-C.

As an example, the various envelope tracking modes may include one ormore envelope tracking power modes of operation including the high powermodulation mode and the medium power modulation mode. As anotherexample, the various average power tracking modes may include one ormore average power tracking modes of operation including a medium poweraverage power tracking mode and a low power average power tracking mode.In one embodiment of the pseudo-envelope tracking modulated power supplysystem 868, the switch mode power supply converter 872 is arranged tocooperatively operate with the parallel amplifier circuit 874 to formthe power management system 870, which operates in one of the high powermodulation mode, the medium power modulation mode, and the low poweraverage power tracking mode. In an alternate embodiment of the powermanagement system 870, the power management system 870 operates in oneof the high power modulation mode, the medium power modulation mode, themedium power average power tracking mode, and the low power averagepower tracking mode. The power management system 870 provides envelopetracking using the power amplifier supply voltage, V_(CC), during thehigh power modulation mode and the medium power modulation mode. Assuch, during the high power modulation mode and the medium powermodulation mode, the power management system 870 controls the poweramplifier supply voltage, V_(CC), to the linear radio frequency poweramplifier 22 to provide the envelope tracking. Further, thepseudo-envelope tracking modulated power supply system 868 providesaverage power tracking during the low power average power tracking mode.As such, during the low power average power tracking mode, the powermanagement system 870 controls the power amplifier supply voltage,V_(CC), to the linear radio frequency power amplifier 22 to provide theaverage power tracking.

In one embodiment of the pseudo-envelope tracking modulated power supplysystem 868, the linear radio frequency power amplifier 22 sequentiallytransmits multiple communications slots. As such, selection of the oneof the high power modulation mode, the medium power modulation mode, andthe low power average power tracking mode is based on an expected outputpower from the radio frequency power amplifier 869 and is made on acommunications slot to communications slot basis. In one embodiment ofthe pseudo-envelope tracking modulated power supply system 868, duringthe low power average power tracking mode, adjustment of a magnitude ofthe power amplifier supply voltage, V_(CC), is made on a communicationsslot to communications slot basis.

The controller 50 may configure the switch mode power supply converter872 to operate in the medium power average power tracking mode when thepower management system 870 is configured to operate in a medium poweraverage power tracking modulation mode. The controller 50 may configurethe switch mode power supply converter 872 to be in a high powermodulation mode when the power management system 870 is configured tooperate in the high power modulation mode. The controller 50 mayconfigure the switch mode power supply converter 872 to be in a mediumpower modulation mode when the power management system 870 is configuredto operate in the medium power modulation mode. The controller 50 mayconfigure the switch mode power supply converter 872 to be in an offmode when the power management system 870 is configured to operate ineither a low power modulation mode or a low power average power trackingmode.

The switch mode power supply converter 872 may include embodiments of aswitcher control circuit 880, a multi-level charge pump 882, a switchingcircuit 884, and an average frequency control circuit 885. The switchercontrol circuit 880, the multi-level charge pump 882, the switchingcircuit 884, and a feedback delay compensation circuit 852 may beconfigured to receive the battery voltage, V_(BAT). Some embodiments ofthe switch mode power supply converter 872 may further include thefeedback delay compensation circuit 852. The controller 50 may configurethe switcher control circuit 880 to govern the operation of themulti-level charge pump 882 and the switching circuit 884 as a functionof the power level modulation mode and the expected output power to begenerated by the embodiments of the radio frequency power amplifier 869in response to a modulated radio frequency input signal to betransmitted. In some embodiments, the switcher control circuit 880 mayalso be configured to control the operation of the feedback delaycompensation circuit 852 as a function of the power level modulationmode and the expected output power of the radio frequency poweramplifier 869. In addition, in some embodiments, the feedback delaycompensation circuit 852 may generate a feedback delay compensationsignal 854, I_(FEEDBACK) _(—) _(TC), as a function of the power levelmodulation mode and the expected output power of the radio frequencypower amplifier 869.

As a non-limiting example, in the high power modulation mode, thecontroller 50 or the switcher control circuit 880 may configure thefeedback delay compensation circuit 852 to operate as a function of ahigh power mode apparent gain to increase the aggressiveness of thefeedback compensation provided by the feedback delay compensation signal854, I_(FEEDBACK) _(—) _(TC). As the apparent gain of the feedback delaycompensation circuit 852 is increased, the switch mode power supplyconverter 872 may respond to a change in the target voltage for themodulated power supply voltage, V_(CC), provided by the differentialV_(RAMP) signal, which may increase the power efficiency of the variousembodiments of a parallel amplifier 928, depicted in FIG. 2C.

In the medium power modulation mode, the controller 50 or the switchercontrol circuit 880 may configure the feedback delay compensationcircuit 852 to operate as a function of a medium power mode apparentgain to decrease the aggressiveness of the feedback compensationprovided by the feedback delay compensation signal 854, I_(FEEDBACK)_(—) _(TC), in order to prevent the switcher control circuit 880 frompre-maturely changing the switching voltage, V_(SW). The feedback delaycompensation circuit 852 may operate in an overly aggressive manner whenthe apparent gain is set too high with respect to the expected outputpower to be generated by the radio frequency power amplifier 869 duringa data transmission. Over aggressiveness of feedback compensationprovided by the feedback delay compensation circuit 852 may result inpre-mature changes in the switching voltage, V_(SW), because the switchmode power supply converter 872 may overreact to a change in the targetvoltage for the modulated power supply voltage, V_(CC), provided by thedifferential V_(RAMP) signal. As a result, the switch mode power supplyconverter 872 may generate a switching voltage that provides too much ortoo little energy to the modulated power supply voltage V_(CC), whichmay decrease the power efficiency of the various embodiments of theparallel amplifier 928, depicted in FIG. 2C.

In some embodiments of the medium power modulation mode, the controller50 may set the apparent gain of the feedback delay compensation circuit852 based on a sliding scale as a function of the expected output powerto be generated by the radio frequency power amplifier 869 during thedata transmission. As an alternative example, some embodiments of theswitcher control circuit 880 may be further configured to adjust theapparent gain of the feedback delay compensation circuit 852 based on asegmentation mode of the switching circuit 884. In some embodiments, thecontroller 50 or the switcher control circuit 880 may disable thefeedback delay compensation circuit 852 during the low power averagepower tracking mode and during the off mode. Some embodiments of thepower management system 870 may enable the feedback delay compensationcircuit 852 in a high power modulation mode and a medium powermodulation mode.

In the low power average power tracking mode, the switcher controlcircuit 880 may disable the multi-level charge pump 882 and theswitching circuit 884, and configure the switching voltage output 26 toprovide a high impedance path to ground. In the low power average powertracking mode, the switch mode power supply converter 872 may beconfigured to disable a clocking signal that may be used by a μC chargepump circuit 262.

In a buck converter “bang-bang” mode of operation, the switcher controlcircuit 880 may disable the multi-level charge pump 882, and control theswitching circuit 884 to swing the switching voltage, V_(SW), betweenthe battery voltage, V_(BAT), and ground. In the multi-level charge pumpbuck converter mode, the switcher control circuit 880 may enable boththe multi-level charge pump 882 and the switching circuit 884. Theswitcher control circuit 880 may further control the multi-level chargepump 882 and the switching circuit 884 to generate both buck levelvoltages and boost level voltages to provide the switching voltage,V_(SW), at the switching voltage output 26 as a function of theoperational state of the switcher control circuit 880 and the powerlevel modulation mode.

In a medium power average power tracking mode of operation, the switchercontrol circuit 880 may be configured to operate in a number of averagepower tracking “bang-bang” modes of operation. As an example, in someembodiments of an average power tracking “bang-bang” mode of operation,the switcher control circuit 880 may configure the multi-level chargepump 882 and the switching circuit 884 to switch the switching voltage,V_(SW), between only a first bang-bang switching voltage, V_(SW) _(—)_(BB1), and a second bang-bang switching voltage, V_(SW) _(—) _(BB2),during a data transmission, where the first bang-bang switching voltage,V_(SW) _(—) _(BB1), is less than the second bang-bang switching voltage,V_(SW) _(—) _(BB2).

Unlike a buck converter mode of operation in which the switching voltageV_(SW) swings between ground and the battery supply, V_(BAT), the firstbang-bang switching voltage, V_(SW) _(—) _(BB1), may be substantiallyequal to ground, the battery voltage, V_(BAT), or the switching voltageV_(SW) between ground and the battery voltage, V_(BAT). The secondbang-bang switching voltage, V_(SW) _(—) _(BB2), may be substantiallyequal to the supply voltage, V_(BAT), or a charge pump output voltagegenerated by the multi-level charge pump 882 from the battery voltage,V_(BAT). The average power tracking “bang-bang” modes of operation mayinclude an average power tracking “buck-buck” mode of operation and anaverage power tracking “buck-boost” mode of operation.

In the average power tracking “buck-buck” mode of operation, theswitcher control circuit 880 may configure the multi-level charge pump882 and the switching circuit 884 to generate a first bang-bangswitching voltage, V_(SW) _(—) _(BB1), and a second bang-bang switchingvoltage, V_(SW) _(—) _(BB2), that are no greater than the batteryvoltage, V_(BAT). For example, the switcher control circuit 880 mayconfigure the multi-level charge pump 882 to generate only a buckedoutput voltage at a charge pump output 64. As an example, the switchercontrol circuit may configure the multi-level charge pump 882 togenerate a first buck mode output voltage, V_(FIRST) _(—) _(BUCK),substantially equal to ½×V_(BAT) in a first buck mode of operation. Inthe average power tracking “buck-buck” mode, the switcher controlcircuit 880 may disable the multi-level charge pump 882 provided thefirst bang-bang switching voltage, V_(SW) _(—) _(BB1), and the secondbang-bang switching voltage, V_(SW) _(—) _(BB2), are a shunt mode outputvoltage substantially equal to ground and a series mode output voltagesubstantially equal to V_(BAT), respectively.

In the average power tracking “buck-boost” mode, the first bang-bangswitching voltage, V_(SW) _(—) _(BB1), may be no greater than thebattery voltage, V_(BAT), and the second bang-bang switching voltage,V_(SW) _(—) _(BB2), is a boost voltage that is greater than the batteryvoltage, V_(BAT). In the average power tracking “buck-boost” mode, theswitcher control circuit 880 may configure the multi-level charge pump882 and the switching circuit 884 to generate the first bang-bangswitching voltage, V_(SW) _(—) _(BB1). The switcher control circuit 880may configure the multi-level charge pump 882 to generate the secondbang-bang switching voltage, V_(SW) _(—) _(BB2). A further descriptionof the average power tracking “bang-bang” modes of operation is providedbelow.

Some embodiments of the switcher control circuit 880 may be configuredto form a composite control signal as a function of the various envelopetracking modes, the various average power tracking modes, and buckconverter modes. As an example, the various embodiments of the switchercontrol circuit 880 may combine the various fractional amounts andcombinations of the feedback delay compensation signal 854, I_(FEEDBACK)_(—) _(TC), the switch mode power supply feedback signal 40A, and thethreshold offset current 42, I_(THRESHOLD) _(—) _(OFFSET), to form oneor more composite control signals as a function of the power levelmodulation mode. In some embodiments, the composite control signal maydepend on the power level modulation mode.

As an example, FIG. 2C depicts an embodiment of the power managementsystem 870 in which a parallel amplifier 928 may generate the switchmode power supply feedback signal 40A in the high power modulation modeand medium power modulation mode. In addition, a V_(OFFSET) loop circuit41 may generate the threshold offset current 42, I_(THRESHOLD) _(—)_(OFFSET), in the high power modulation mode and the medium powermodulation mode.

Returning to FIG. 2B, the switcher control circuit 880 may also providea series switch control signal 66 and a shunt switch control signal 68to the switching circuit 884. In response to an assertion of the seriesswitch control signal 66, the switching circuit 884 couples theswitching voltage output 26 to the battery voltage, V_(BAT), to generatethe switching voltage, V_(SW), substantially equal to V_(BAT). Inresponse to the shunt switch control signal 68, the switching circuit884 couples the switching voltage output 26 to ground to generate theswitching voltage, V_(SW), substantially equal to zero. In addition, theswitcher control circuit 880 provides a segmentation control signal 880Ato the switching circuit 884 as a function of the power level modulationmode. In some embodiments, the switcher control circuit 880 may generatethe segmentation control signal 880A as a function of the power levelmodulation mode and the expected output power to be generated by theradio frequency power amplifier 869 during a data transmission. Forexample, some embodiments of the power management system 870 may beconfigured to generate the segmentation control signal 880A based on anaverage expected output power, P_(OUT) _(—) _(AVE), of the radiofrequency power amplifier 869 that maximizes the efficiency of theswitch mode power supply converter 872.

In general, the switch mode power supply converter 872 has a segmentedoutput stage, such that during the high power modulation mode and themedium power modulation mode, segment selection of the segmented outputstage is based on an expected output power from the radio frequencypower amplifier 869 to increase efficiency of the pseudo-envelopetracking modulated power supply system 868.

The switcher control circuit 880 may configure the switch mode powersupply converter 872 to generate a switching voltage, V_(SW), at theswitching voltage output 26 based on an operational state of theswitcher control circuit 880 as a function of a power level modulationmode, which is dependent on the expected output power to be generated bythe radio frequency power amplifier 869 during a data transmission. Insome embodiments, the switcher control circuit 880 may configure themulti-level charge pump 882 and the switching circuit 884 to operate inone of a buck converter “bang-bang” mode, a multi-level charge pump buckconverter mode, and an average power tracking mode of operation as afunction of the power level modulation mode and operational mode of theswitcher control circuit 880.

Referring to FIGS. 2C and 2B, in some embodiments, a control signal 926,received at the non-inverting input of the parallel amplifier 928 togenerate the parallel amplifier output current, I_(PARA) _(—) _(AMP),may be generated by a differential filter 924 as a function of the powerlevel modulation mode. For example, in the high power modulation mode,the differential filter 924 may provide an increased level of frequencycompensation or correction as compared to the frequency compensation orcorrection provided by the differential filter 924 in the medium powermodulation mode. Accordingly, some embodiments of the switcher controlcircuit 880, the switch mode power supply converter 872, the parallelamplifier circuit 874 and/or a combination thereof, may be furtheradapted to adjust a scaling factor, M, provided to a buffer scalar 434,and/or the magnitude of the delayed I_(COR) estimated switching voltageoutput 38C, V_(SW) _(—) _(EST) _(—) _(DELAY) _(—) _(ICOR), as a functionof the power level modulation mode.

Some alternative embodiments of the switcher control circuit 880, theswitch mode power supply converter 872, the parallel amplifier circuit874, and/or combinations thereof, may be further configured to adjustthe magnitude of the delayed I_(COR) estimated switching voltage output38C, V_(SW) _(—) _(EST) _(—) _(DELAY) _(—) _(ICOR), based on themagnitude of the control signal 926 that is received by at thenon-inverting input of the parallel amplifier 928, which is generated bythe differential filter 924. In some embodiments, the switcher controlcircuit 880 may selectively adjust the scaling factor, M, as a functionof the magnitude of the differential V_(RAMP) signal and the power levelmodulation mode. For example, in some embodiments of the switchercontrol circuit 880, the scaling factor, M, may be further adjusted as afunction of the power level modulation mode to reflect the amount offrequency compensation or correction applied by the differential filter924 to generate the control signal 926 received at the non-invertinginput 928A of the parallel amplifier 928, depicted in FIG. 2C.

In some embodiments of the power management system 870, either theswitch mode power supply converter 872 or the parallel amplifier circuit874 may be further adapted to adjust the scaling factor, M, based on themagnitude of the control signal 926 or the power level modulation mode.For example, in some embodiments, the magnitude of the delayed I_(COR)estimated switching voltage output 38C, V_(SW) _(—) _(EST) _(—) _(DELAY)_(—) _(ICOR), may be adjusted as a function of the power levelmodulation mode, the magnitude of the differential V_(RAMP) signal, andthe expected frequency response of the parallel amplifier 928 whenoperating in each of the respective power level modulation modes.

For example, a switcher propagation delay is a time period between whenthe state of the switcher control circuit 880 changes to a newoperational state and the switching voltage, V_(SW), generated inresponse to the new operational state of the switcher control circuit880, is generated at the switching voltage output 26. In someembodiments of the switch mode power supply converter 872, the switcherpropagation delay may vary based on being configured to operate in theenvelope tracking mode of operation and the average power tracking mode.Thus, the controller 50 may be configured to adjust the programmabledelay period as a function of an expected output power to be generatedby the radio frequency power.

As another example, in some embodiments, the switching circuit 884 maybe a segmented switching circuit having a number of sets of seriesswitch and shunt switch pairs. The sets of series switch and shuntswitch pairs may be arranged in parallel and operably coupled to form asegmented series switch and shunt switch network. The switcher controlcircuit 880 may generate a series switch control signal 66, a shuntswitch control signal 68, and the segmentation control signal 880A basedon an operational state of the switcher control circuit 880 to controlthe operation of the switching circuit 884. In addition, thesegmentation control signal 880A may be dependent on the power levelmodulation mode of the power management system 870. For example, theswitcher control circuit 880 may configure the segmentation controlsignal 880A to enable or disable various combinations of the sets ofseries switch and shunt switch pairs of the switching circuit 884 basedon the power level modulation mode of operation of the respective powermanagement system 870, depicted in FIG. 2A-C. As an example, in someembodiments of the medium power modulation mode, the switcher controlcircuit 880 may generate a segmentation control signal 880A to enableeither 75% or 50% of the number of sets of series switch and shuntswitch pairs of the switching circuit 884.

In some embodiments of the switching circuit 884, the switcherpropagation delay may vary depending on the number of sets of seriesswitch and shunt switch pairs configured to operate during a datatransmission. In some embodiments, the switcher control circuit 880 maybe configured to adjust a programmable delay period depending on thesegmentation control signal 880A to maintain the temporal alignment ofthe delayed I_(COR) estimated switching voltage output 38C V_(SW) _(—)_(EST) _(—) _(DELAY) _(—) _(ICOR) with respect to the switching voltage,V_(SW), provided at the switching voltage output 26.

The switching circuit 884 may include a segmentation control circuit 886configured to receive a segmentation control signal 880A from theswitcher control circuit 880. The switching circuit 884 may furtherinclude segmented series switches 888 and segmented shunt switches 890in communication with the segmentation control circuit 886. Thesegmented series switches 888 may include a first series switch 892, asecond series switch 894, a third series switch 896, and a fourth seriesswitch 898. The segmented shunt switches 890 may include a first shuntswitch 900, a second shunt switch 902, a third shunt switch 904, and afourth shunt switch 906.

A source 892S of the first series switch 892, a source 894S of thesecond series switch 894, a source 896S of the third series switch 896,and a source 898S of the fourth series switch 898 are in communicationwith a supply voltage received from the battery 20 substantially equalto the battery voltage, V_(BAT). A drain 892D of the first series switch892, a drain 894D of the second series switch 894, a drain 896D of thethird series switch 896, and a drain 898D of the fourth series switch898 are respectively coupled to a drain 900D of the first shunt switch900, a drain 902D of the second shunt switch 902, a drain 904D of thethird shunt switch 904, and a drain 906D of the fourth shunt switch 906to form the switching voltage output 26. A source 900S of the firstshunt switch 900, a source 902S of the second shunt switch 902, a source904S of the third shunt switch 904, and a source 906S of the fourthshunt switch 906 are in communication with a reference voltagesubstantially equal to ground.

The segmentation control circuit 886 may include a first series switchcontrol output 908 coupled to the gate 892G of the first series switch892, a first shunt switch control output 910 coupled to the gate 900G ofthe first shunt switch 900, a second series switch control output 912coupled to the gate 894G of the second series switch 894, a second shuntswitch control output 914 coupled to the gate 902G of the second shuntswitch 902, a third series switch control output 916 coupled to the gate896G of the third series switch 896, a third shunt control output 918coupled to the gate 904G of the third shunt switch 904, a fourth seriesswitch control output 920 couple to the gate 898G of the fourth seriesswitch 898, and a fourth shunt switch control output 922 coupled to thegate 906G of the fourth shunt switch 906.

Operationally, the segmentation control circuit 886 may divide thesegmented series switches 888 and segmented shunt switches 890 intological groupings of sets of series and shunt switches such that eachset of series and shunt switches includes one of the segmented seriesswitches 888 and one of the segmented shunt switches 890. For example, afirst set of series and shunt switches may include the first seriesswitch 892 and the first shunt switch 900. A second set of series andshunt switches may include the second series switch 894 and the secondshunt switch 902. A third set of series and shunt switches may includethe third series switch 896 and the third shunt switch 904. A fourth setof series and shunt switches may include the fourth series switch 898and the fourth shunt switch 906.

The segmentation control circuit 886 is configured to receive a seriesswitch control signal 66 and a shunt switch control signal 68 from theswitcher control circuit 880. The power management system 870 mayconfigure the switcher control circuit 880 to generate a segmentationcontrol signal 880A to select which of the segmented series switches 888and segmented shunt switches 890 are to be enabled by the segmentationcontrol circuit 886 to generate the switching voltage, V_(SW), at theswitching voltage output 26 as a function of the average expected outputpower, P_(OUT) _(—) _(AVE), to be generated by the radio frequency poweramplifier 869 during the data transmission. Illustratively, in the casewhere there are four sets of series and shunt switches, the operation ofthe segmentation control circuit 886 may be divided into four regions ormodes of operation.

In some embodiments, the switcher control circuit 880 may generate thesegmentation control signal 880A as a function of the average expectedoutput power, P_(OUT) _(—) _(AVE), and a set of switcher output powerthresholds including a first switcher output power threshold, P_(OUT1),a second switcher output power threshold, P_(OUT2), and a third switcheroutput power threshold, P_(OUT3). The values of the first switcheroutput power threshold, P_(OUT1), the second switcher output powerthreshold, P_(OUT2), and the third switcher output power threshold,P_(OUT3), may be determined to maximize the efficiency of the switchmode power supply converter 872 with respect to the expected outputpower of the radio frequency power amplifier 869 as a function of thepower level modulation mode and an expected load line, R_(RF) _(—)_(AMP) _(—) _(EXP), of the radio frequency power amplifier 869 duringthe data transmission.

In the case where the average expected output power, P_(OUT) _(—)_(AVE), of the radio frequency power amplifier 869 is at least equal tothe first switcher output power threshold, P_(OUT1), the segmentationcontrol signal 880A may configure the segmentation control circuit 886to enable all four of the series switch and shunt switch segments. Inthe case where the average expected output power, P_(OUT) _(—) _(AVE),of the radio frequency power amplifier 869 is less than the firstswitcher output power threshold, P_(OUT1), and at least equal to thesecond switcher output power threshold, P_(OUT2), the segmentationcontrol signal 880A may configure the segmentation control circuit 886to enable three of the series switch and shunt switch segments. In thecase where the average expected output power, P_(OUT) _(—) _(AVE), ofthe radio frequency power amplifier 869 is less than the second switcheroutput power threshold, P_(OUT2), and at least equal to the thirdswitcher output power threshold, P_(OUT3), the segmentation controlsignal 880A may configure the segmentation control circuit 886 to enabletwo of the series switch and shunt switch segments. And in the casewhere the average expected output power, P_(OUT) _(—) _(AVE), of theradio frequency power amplifier 869 is less than the third switcheroutput power threshold, P_(OUT3), the segmentation control signal 880Amay configure the segmentation control circuit 886 to enable one of theseries switch and shunt switch segments.

The multi-level charge pump 882 may configure a multi-level charge pumpswitching network to generate a variety of “boost” output voltages and“buck” output voltages as a function of a charge pump mode controlsignal 62 received from the switcher control circuit 880. Someembodiments of the multi-level charge pump 882 may generate a variety of“boost” output voltages and “buck” output voltages as a multi-levelcharge pump output voltage, V_(MLCP), in response to the charge pumpmode control input 62 generated by the switcher control circuit 880. Themulti-level charge pump 882 may provide the multi-level charge pumpoutput voltage, V_(MLCP), to the charge pump output 64, which may becoupled through the switching circuit 884 to the switching voltageoutput 26.

For example, in a second boost mode of operation, the multi-level chargepump 882 may configure the multi-level charge pump switching network togenerate a second boost mode output voltage, V_(SECOND) _(—) _(BOOST),substantially equal to 2×V_(BAT) at the charge pump output 64. In afirst boost mode of operation, the multi-level charge pump 882 mayconfigure the switching network to generate a first boost mode outputvoltage at the charge pump output 64 substantially equal to 3/2×V_(BAT).In a first buck mode of operation, the multi-level charge pump 882 mayconfigure the multi-level charge pump switching network to generate afirst buck mode output voltage at the charge pump output 64substantially equal to ½×V_(BAT). In some alternative embodiments, themulti-level charge pump 882 may be configured to generate other ratiosof boost mode output voltages and buck mode output voltage.

Some embodiments of the multi-level charge pump 882 may include only twoflying capacitors that are coupled in various switching arrangementswith respect to each other, the battery voltage, V_(BAT), and the chargepump output 64 to generate the various charge pump output voltages atthe charge pump output 64. For example, similar to the μC charge pumpcircuit 262, some embodiments of the multi-level charge pump 882 maygenerate various ratios of output voltages that may be provided as aboost output voltage or a buck output voltage to the switching voltageoutput 26 as the switching voltage, V_(SW). In still other alternativeembodiments, the multi-level charge pump 882 may generate a boost outputvoltage or a buck output voltage with respect to a supply voltage otherthan the battery voltage, V_(BAT), where the supply voltage may begreater than the battery voltage, V_(BAT), or less than the batteryvoltage, V_(BAT). In some alternative embodiments, the supply voltageprovided to the multi-level charge pump 882 may be a boosted voltage ora bucked voltage derived from the battery voltage, V_(BAT).

Based on the power level modulation mode, in some embodiments of themulti-level charge pump buck converter mode of operation, the switchercontrol circuit 880 may configure the switch mode power supply converter872 to generate a series output voltage substantially equal to thebattery voltage, V_(BAT), a shunt output voltage substantially equal toa ground voltage, and a subset of the available charge pump outputvoltages. As an example, as a function of the power level modulationmode, the switcher control circuit 880 may configure the multi-levelcharge pump 882 and the switching circuit 884 to generate a shunt modeoutput voltage substantially equal to a ground voltage in the shuntoutput mode, a series mode output voltage substantially equal to V_(BAT)in the series output mode, and a first boost mode output voltageV_(FIRST) _(—) _(BOOST) substantially equal to 3/2×V_(BAT) in the firstboost output mode. As another non-limiting example, as a function of thepower level modulation mode, in some power level modulation modes, theswitcher control circuit 880 may configure the multi-level charge pump882 to provide a first buck mode output voltage substantially equal to½×V_(BAT) in the first buck mode of operation. As an example, as afunction of the power level modulation mode, the switcher controlcircuit 880 may configure the multi-level charge pump 882 and theswitching circuit 884 to generate a shunt mode output voltagesubstantially equal to ground, a first buck mode output voltagesubstantially equal to ½×V_(BAT), a series mode output voltagesubstantially equal to V_(BAT), and a first boost mode output voltagesubstantially equal to 3/2×V_(BAT) as a function of the operationalstate of the switcher control circuit 880. As still another non-limitingexample, in some power level modulation modes, the switcher controlcircuit 880 may configure the multi-level charge pump 882 to onlyoperate in a first boost mode of operation to generate a first boostmode output voltage, V_(FIRST) _(—) _(BOOST), substantially equal to3/2×V_(BAT).

As another example, in some power level modulation modes, themulti-level charge pump 882 may be configured to only operate in thesecond boost mode of operation. In still another example, in some powerlevel modulation modes, the multi-level charge pump 882 may beconfigured to operate in the first buck mode and either the first boostmode or the second boost mode of operation. For example, in some powerlevel modulation modes, the switcher control circuit 880 may configurethe multi-level charge pump 882 to provide either a first buck modeoutput voltage substantially equal to ½×V_(BAT) and a first boost modeoutput voltage substantially equal to 3/2×V_(BAT) as a function of theoperational state of the switcher control circuit 880. In other powerlevel modulation modes, the switcher control circuit may configure themulti-level charge pump 882 to provide a buck output voltagesubstantially equal to ½×V_(BAT) in the first buck mode and a secondboost level output voltage substantially equal to 2×V_(BAT) as afunction of the operational state of the switcher control circuit 880.

Depending on a power level modulation mode, a characteristic of the datatransmission, and/or a desired distribution of harmonics of theswitching frequency within the frequency spectrum, the controller 50 mayconfigure the comparator thresholds to set the equivalent main ripplevoltage level at the modulated power supply output 876 and/or theswitching frequency of the switch mode power supply converter 872. Forexample, in the medium power average power tracking modulation mode

The parallel amplifier 928 may include a non-inverting input 928Aconfigured to receive the control signal 926. In some embodiments, thecontrol signal 926 may be a compensated V_(RAMP) signal, V_(RAMP) _(—)_(C). The parallel amplifier 928 may also include an inverting input928B in communication with the modulated power supply output 876. Basedon the difference between the control signal and the modulated powersupply voltage, V_(CC), the parallel amplifier 928 may generate aparallel amplifier output current, I_(PARA) _(—) _(AMP), at the parallelamplifier output 928C in order to generate the parallel amplifier outputvoltage, V_(PARA) _(—) _(AMP) at the parallel amplifier circuit output874A during the high power modulation mode and the medium powermodulation mode. Additionally, the parallel amplifier 928 may bedisabled during the low power average power tracking mode. In anotherembodiment of the parallel amplifier 928, the parallel amplifier 928 isdisabled during the medium power average power tracking mode.

The parallel amplifier 928 may generate a scaled parallel amplifieroutput current estimate, I_(PARA) _(—) _(AMP) _(—) _(SENSE), which is afractional representation of the parallel amplifier output current,I_(PARA) _(—) _(AMP), from the parallel amplifier feedback output 928E.The parallel amplifier 928 may include a parallel amplifier supplyvoltage input 928D configured to receive a μC charge pump outputvoltage, V_(μC) _(—) _(OUT), from the μC charge pump circuit 262, suchthat the parallel amplifier output voltage, V_(PARA) _(—) _(AMP) isbased on the μC charge pump output voltage, V_(μC) _(—) _(OUT) duringthe high power modulation mode and the medium power modulation mode. Ingeneral, the power amplifier supply voltage, V_(CC), is based on the μCcharge pump output voltage, V_(μC) _(—) _(OUT), during the high powermodulation mode and the medium power modulation mode.

In one embodiment of the parallel amplifier 928, the parallel amplifier928 has a segmented output stage, such that during the high powermodulation mode and the medium power modulation mode, segment selectionof the segmented output stage is based on an expected output power fromthe radio frequency power amplifier 869 to increase efficiency of thepseudo-envelope tracking modulated power supply system 868.

Alternatively, as previously described, in some embodiments the parallelamplifier supply voltage input 928D may be switchably configured to bepowered by the μC charge pump output voltage, V_(μC) _(—) _(OUT), or asupply voltage provided by the multi-level charge pump 882 of the switchmode power supply converter 872, depicted in FIG. 2B.

For example, in some embodiments, the μC charge pump circuit 262 maygenerate the μC charge pump output voltage, V_(μC) _(—) _(OUT), as afunction of the battery voltage and the modulation mode of operation. Assuch, the μC charge pump output voltage, V_(μC) _(—) _(OUT), may bebased on a charge-pump-ratio, which may be based on one of at least thehigh power modulation mode, the medium power modulation mode, and thelow power average power tracking mode. For example, in the high powermodulation mode, the power management system 870 may configure the μCcharge pump circuit 262 to operate in the 1×V_(BAT) mode or the4/3×V_(BAT) mode to generate the μC charge pump output voltage, V_(μC)_(—) _(OUT), substantially equal to either the battery voltage,1×V_(BAT), or 4/3×V_(BAT), respectively, which equates to acharge-pump-ratio of 1 or 4/3, respectively. In the medium powermodulation mode, the power management system 870 may configure the μCcharge pump circuit 262 to operate in the 1×V_(BAT) mode or the⅔×V_(BAT) mode to generate the μC charge pump output voltage, V_(μC)_(—) _(OUT), substantially equal to either the battery voltage, V_(BAT),or ⅔×V_(BAT), respectively, which equates to a charge-pump-ratio of 1 or⅔, respectively. In some embodiments, in the low power modulation mode,the power management system 870 may configure the μC charge pump circuit262 to operate in the ¼×V_(BAT) mode, ⅓×V_(BAT) mode, or the ½×V_(BAT)mode to generate the μC charge pump output voltage, V_(μC) _(—) _(OUT),substantially equal to ¼×V_(BAT), ⅓×V_(BAT), or ½×V_(BAT), respectively,which equates to a charge-pump-ratio of ¼, ⅓, or ½, respectively.

In one embodiment of the μC charge pump circuit 262, the μC charge pumpcircuit 262 provides the μC charge pump output voltage, V_(μC) _(—)_(OUT) based on the battery voltage, V_(BAT). In one embodiment of theμC charge pump circuit 262, the μC charge pump circuit 262 is acapacitor-based charge pump circuit 262, such that multiple switchedflying capacitors are used to provide the μC charge pump output voltage,V_(μC) _(—) _(OUT) In an alternate embodiment of the parallel amplifiercircuit 874, the μC charge pump circuit 262 is replaced with a μL chargepump circuit (not shown), which provides the μC charge pump outputvoltage, V_(μC) _(—) _(OUT) based on the battery voltage, V_(BAT). TheμL charge pump circuit (not shown) is an inductor-based charge pumpcircuit, such that at least one inductor is used to provide the μCcharge pump output voltage, V_(μC) _(—) _(OUT) As such, either the μCcharge pump circuit 262, the μL charge pump circuit (not shown), or bothmay operate in a boost, a buck mode, or both. As such, the μC chargepump output voltage, V_(μC) _(—) _(OUT) may be greater than, equal to,or less than the battery voltage, V_(BAT).

In one embodiment of the μC charge pump circuit 262, the μC charge pumpcircuit 262 has a segmented output stage, such that during the highpower modulation mode, during the medium power modulation mode, andduring the low power average power tracking mode, segment selection ofthe segmented output stage is based on an expected output power from theradio frequency power amplifier 869 to increase efficiency of thepseudo-envelope tracking modulated power supply system 868.

In some embodiments, a segmented parallel amplifier output stage of theparallel amplifier 928 may be configured based upon the expected outputpower of the radio frequency power amplifier 869. As an example, thesegmentation configuration of the parallel amplifier 928 may be afunction of a maximum instantaneous output power, P_(INST) _(—) _(MAX),to be generated by the radio frequency power amplifier 869 during thedata transmission and the expected load line, R_(RF) _(—) _(AMP) _(—)_(EXP), of the radio frequency power amplifier 869 during the datatransmission. As an example, in some embodiments, a maximuminstantaneous output current, I_(MAX) _(—) _(PARA) _(—) _(AMP), to beprovided by the parallel amplifier 928 during the data transmission maybe substantially equal to (P_(INST) _(—) _(MAX)/R_(RF) _(—) _(AMP) _(—)_(EXP))^(1/2). In some embodiments, the parallel amplifier 928 mayinclude a maximum parallel amplifier output current, I_(PARA) _(—)_(MAX), which is the maximum output current to be generated by theparallel amplifier 928. In some embodiments, the power management system870 may configure the segmentation configuration of the parallelamplifier 928 based on the maximum parallel amplifier output current,I_(PARA) _(—) _(MAX), and the maximum instantaneous output current,I_(MAX) _(—) _(PARA) _(—) _(AMP).

For example, in some embodiments, the controller may determine themaximum instantaneous output current, I_(MAX) _(—) _(PARA) _(—) _(AMP),based on the envelope characteristics of the modulated radio frequencyinput signal to be transmitted and the expected load line, R_(RF) _(—)_(AMP) _(—) _(EXP), of the radio frequency power amplifier 869 duringthe data transmission. The power management system 870 may determine themaximum instantaneous output power, P_(INST) _(—) _(MAX), based on theenvelope characteristics of the modulated radio frequency input signal.Based on the maximum instantaneous output current, I_(MAX) _(—) _(PARA)_(—) _(AMP), the power management system 870 may determine an estimatedpercentage of output current generation capacity of the parallelamplifier 928 that may be used during the data transmission.

For example, the power management system 870 may calculate thepercentage of the output current generation capacity based on the ratioof the maximum instantaneous output current, I_(MAX) _(—) _(PARA) _(—)_(AMP), to the maximum parallel amplifier output current, I_(PARA) _(—)_(MAX). The power management system 870 may determine the number ofoutput stage segments of the parallel amplifier 928 to enable based onthe estimated percentage of output current generation capacity of theparallel amplifier 928 to be used. For example, in the case where theparallel amplifier 928 includes two output stage segments that areconfigured to have a substantially equal output current generationcapacity, the power management system 870 may set the segmentationconfiguration to be 100% when the ratio of the maximum instantaneousoutput current, I_(MAX) _(—) _(PARA) _(—) _(AMP), to the maximumparallel amplifier output current, I_(PARA) _(—) _(MAX), issubstantially equal to or near at least 50%. The power management system870 may set the segmentation configuration to be 50% when the ratio ofthe maximum instantaneous output current, I_(MAX) _(—) _(PARA) _(—)_(AMP), to the maximum parallel amplifier output current, I_(PARA) _(—)_(MAX), is at least less than 50%. In the case where the output stagesegments of the parallel amplifier 928 are not substantially equal, thecontroller 50 may determine which of the output stage segments to enablebased on the ratio of the maximum instantaneous output current, I_(MAX)_(—) _(PARA) _(—) _(AMP), and the current generation capacity of each ofthe output stage segments.

In some alternative embodiments, the segmentation configuration of theparallel amplifier 928 may be based on the expected peak-to-peak swingof a modulated power supply voltage, V_(CC) _(—) _(PKPK), and theexpected load line, R_(RF) _(—) _(AMP) _(—) _(EXP), of the radiofrequency power amplifier 869 during the data transmission.

For example, when operating in the high power modulation mode, the powermanagement system 870 may set the segmentation configuration to be 100%.Alternatively, the power management system 870 may configure theparallel amplifier 928 to use only the first output stage segment oronly the second output stage segment while operating in the medium powermodulation mode depending on the ratio of the maximum instantaneousoutput current, I_(MAX) _(—) _(PARA) _(—) _(AMP), to the maximumparallel amplifier output current, I_(PARA) _(—) _(MAX). And in the lowpower modulation mode, the power management system 870 may disable theparallel amplifier 928 to place the parallel amplifier output 928C in ahigh impedance mode.

As previously described with respect to FIG. 2A, FIG. 2C depicts thatthe parallel amplifier circuit 874 may provide the threshold offsetcurrent 42, I_(THRESHOLD) _(—) _(OFFSET), from an embodiment of theV_(OFFSET) loop circuit 41 to regulate the offset voltage, V_(OFFSET),across the coupling capacitor 18A.

In some embodiments, the V_(OFFSET) loop circuit 41 may be configured togenerate the threshold offset current 42, I_(THRESHOLD) _(—) _(OFFSET),that represents an average or integrated error between the modulatedpower supply voltage, V_(CC), and a V_(RAMP) signal when the switch modepower supply converter 872 is configured to operate in an average powertracking mode of operation. Some embodiments of the V_(OFFSET) loopcircuit 41 may be configured to pre-charge the bypass capacitor 19 andthe coupling capacitor 18A while the switch mode power supply converter872 is configured to operate in the average power tracking mode ofoperation.

The parallel amplifier circuit 874 may further include a linearregulator 930, LDO. The linear regulator 930, LDO, may be a low dropoutvoltage linear regulator. The parallel amplifier circuit 874 may alsoinclude a parallel amplifier output bypass switch 936, a linearregulator output selection switch 938, and a feedback selection switch940. The parallel amplifier output bypass switch 936 includes a firstterminal 936A coupled to the parallel amplifier output 928C and a secondterminal 936B coupled to ground. The power management system 870 mayconfigure the parallel amplifier output bypass switch 936 to close whenthe parallel amplifier 928 is disabled.

For example, the power management system 870 may configure the parallelamplifier output bypass switch 936 to close based on a determinationthat the expected output power of the radio frequency power amplifier869 is less than the low power modulation mode threshold or the parallelamplifier output 928C is disabled and configured to provide a highimpedance. Alternatively, the power management system 870 may configurethe parallel amplifier output bypass switch 936 to be open when theparallel amplifier 928 is enabled. As such, the parallel amplifieroutput bypass switch 936 is coupled between the parallel amplifiercircuit output 874A and ground. During the high power modulation modeand the medium power modulation mode, the parallel amplifier outputbypass switch 936 is OPEN and during the low power average powertracking mode, the parallel amplifier output bypass switch 936 isCLOSED.

The linear regulator output selection switch 938 may include an inputterminal 938A coupled to a linear regulator output 930C of the linearregulator 930, LDO, a first output terminal 938B in communication withthe modulated power supply output 876, and a second output terminal 938Cin communication with the second linear regulator output 874D, LDO₂OUTPUT, in communication with the CMOS logic supply input 869C of theradio frequency power amplifier 869. In one embodiment of the powermanagement system 870, the μC charge pump circuit 262 provides thesecond linear regulator output voltage, V_(LDO2), to PA CMOS biascircuitry in the radio frequency power amplifier 869 via the linearregulator 930, LDO.

The feedback selection switch 940 includes an output terminal 940A incommunication with an inverting input 930B of the linear regulator 930,LDO, a first input terminal 940B in communication with the modulatedpower supply output 876, and a second input terminal 940C incommunication with the second linear regulator output 874D, LDO₂ OUTPUT,of the parallel amplifier circuit 874. The linear regulator 930, LDO,also includes a linear regulator power supply input 930D configured toreceive the μC charge pump output voltage, V_(μC) _(—) _(OUT). The μCcharge pump output voltage, V_(μC) _(—) _(OUT), may be configuredprovide a voltage level to the linear regulator power supply input 930Das a function of the battery voltage, V_(BAT), and the expected outputpower of the radio frequency power amplifier 869.

The linear regulator 930, LDO, may also include a non-inverting input930A in communication with the non-inverting input 928A of the parallelamplifier. The non-inverting input 930A of the linear regulator 930,LDO, may also be in communication with the differential filter 924, andconfigured to receive the control signal 926 at the non-inverting input930A. The linear regulator 930, LDO, may also receive a linear regulatorfeedback signal, LDO FEEDBACK, from the output terminal 940A of thefeedback selection switch 940. Based on the difference between thecontrol signal 926 and the linear regulator feedback signal, LDOFEEDBACK, the linear regulator 930, LDO, may generate a linear regulatoroutput voltage at the linear regulator output 930C. Based on the switchstates of the linear regulator output selection switch 938 and thefeedback selection switch 940, the linear regulator 930, LDO, maygenerate a linear regulator output voltage to apply the first linearregulator output current 878A, I_(LDO), to the modulated power supplyoutput 876.

In this regard, in one embodiment of the linear regulator outputselection switch 938 and the linear regulator 930, LDO, during the lowpower average power tracking mode, the linear regulator 930, LDO,provides the power amplifier supply voltage, V_(CC) based on the μCcharge pump output voltage, V_(μC) _(—) _(OUT). As such, the poweramplifier supply voltage, V_(CC), is based on the μC charge pump outputvoltage, V_(μC) _(—) _(OUT), during the low power average power trackingmode. Further, in one embodiment of the linear regulator 930, LDO,during the medium power modulation mode, the linear regulator 930, LDO,is disabled. Additionally, in one embodiment of the linear regulator930, LDO, during the high power modulation mode, the linear regulator930, LDO, is disabled.

In addition, in one embodiment of the linear regulator 930, LDO, duringthe medium power average power tracking mode, the linear regulator 930,LDO, is disabled. In one embodiment of the linear regulator 930, LDO,and the parallel amplifier 928, during the medium power average powertracking mode, both the linear regulator 930, LDO, and the parallelamplifier 928, are disabled. Further, in one embodiment of the switchmode power supply converter 872, during the medium power average powertracking mode, the switch mode power supply converter 872 provides thepower amplifier supply voltage, V_(CC).

FIG. 3 depicts a method 1700 for configuring the operation of the powermanagement system 870 (FIG. 2A). The power management system 870 maydetermine an expected output power to be generated by a radio frequencypower amplifier 869 (FIG. 2A) during a data transmission. (Step 1702).For example, in some embodiments, the controller 50 (FIG. 2A) maydetermine the expected output power to be generated by the radiofrequency power amplifier 869 (FIG. 2A) based upon the envelopecharacteristics of a data signal to be transmitted during a datatransmission slot. In other embodiments, the digital baseband processingportion of the transceiver or modem that provides the differentialV_(RAMP) signal to the parallel amplifier circuit 874 (FIG. 2A) maydetermine the expected output power to be generated by the radiofrequency power amplifier 869 (FIG. 2A).

Based on the expected output power to be generated by the radiofrequency power amplifier 869 (FIG. 2A) during the data transmission,the power management system 870 (FIG. 2A) may select a power modulationmode of operation from among a number of power level modulation modes ofoperation. (Step 1704). For example, the power management system 870(FIG. 2A) includes the high power modulation mode, the medium powermodulation mode, and the low power average power tracking mode. As such,the power management system 870 (FIG. 2A) selects the one of the highpower modulation mode, the medium power modulation mode, and the lowpower average power tracking mode.

In the case where the expected output power to be generated by the radiofrequency power amplifier 869 (FIG. 2A) is greater than or equal to ahigh power modulation mode threshold, the power management system 870(FIG. 2A) configures the power management system 870 (FIG. 2A) tooperate in the high power modulation mode. (Step 1706). Alternatively,in the case where the expected output power to be generated by the radiofrequency power amplifier 869 (FIG. 2A) is less than the high powermodulation mode threshold but greater than or equal to a medium powermodulation mode threshold, the power management system 870 (FIG. 2A)configures the power management system 870 (FIG. 2A) to operate in themedium power modulation mode. (Step 1708).

In the case where the expected output power to be generated by the radiofrequency power amplifier 869 (FIG. 2A) is less than the medium powermodulation mode threshold but greater than or equal to the low powermodulation mode threshold, the power management system 870 (FIG. 2A) mayconfigure the power management system 870 (FIG. 2A) to operate in themedium power average power tracking mode. (Step 1710). In the case wherethe expected output power to be generated by the radio frequency poweramplifier 869 (FIG. 2A) is less than the low power modulation modethreshold, the power management system 870 (FIG. 2A) may configure thepower management system 870 (FIG. 2A) to operate in the low poweraverage power tracking mode. (Step 1712).

In another embodiment of the power management system 870 (FIG. 2A), inthe case where the expected output power to be generated by the radiofrequency power amplifier 869 (FIG. 2A) is less than the medium powermodulation mode threshold, the power management system 870 (FIG. 2A) mayconfigure the power management system 870 (FIG. 2A) to operate in thelow power average power tracking mode (Step 1712).

In one embodiment of the power management system 870 (FIG. 2A), the highpower modulation mode threshold is equal to about three decibels lessthan a maximum expected output power from the radio frequency poweramplifier 869 (FIG. 2A). In one embodiment of the power managementsystem 870 (FIG. 2A), the medium power modulation mode threshold isequal to about ten decibels less than a maximum expected output powerfrom the radio frequency power amplifier 869 (FIG. 2A).

FIG. 5 depicts a portion of a communication device including variousembodiments of an envelope tracking modulated power supply system 2500with transceiver power control loop compensation. For the purposes ofillustration and not by way of limitation, the envelope trackingmodulated power supply system 2500 with transceiver power control loopcompensation may be described with continuing reference to a number ofthe embodiments of the pseudo-envelope tracking modulated power supplysystem 868, depicted in FIGS. 2A-C.

The envelope tracking modulated power supply system 2500 withtransceiver power control loop compensation may include a transceiverpower control loop 2502 and an envelope tracking control system 2503.The transceiver power control loop 2502 may include a transmit-targetpower setting parameter 2502A and a transmit-gain setting parameter2502B. The transceiver power control loop 2502 may adjust a drive levelor an input power P_(IN) of a modulated radio frequency input signal2504 provided to a radio frequency power amplifier 2540 based on thevalues of the transmit-target power setting parameter 2502A and thetransmit-gain setting parameter 2502B as a function of the output powerP_(OUT) generated by the radio frequency power amplifier 2540.

In some embodiments, the modulated radio frequency input signal 2504,P_(IN), may be a test signal. As an example, the modulated radiofrequency input signal 2504, P_(IN), may be a carrier wave having adesired frequency.

In some embodiments of the communication device, the transmit-targetpower setting parameter 2502A may be based on network controlinformation provided by a communication network in which thecommunication device is configured to operate. For example, thecommunication network may provide the communication device with networkcontrol information to configure the operation of the communicationdevice within the communication network. In some embodiments, thetransmit-target power setting parameter 2502A used by a communicationdevice may be a function of the different wireless transmissionsgenerated by the communication device. The transmit-target power settingparameter 2502A of the communication device may also be based on a datarate of the transmission, the resource blocks to be used during thetransmission, a frequency band allocated for the transmission, therelative location of the transmission band to other bands of operationin the communication network, and/or a combination thereof. As anotherexample, the transmit-target power setting parameter 2502A may be afunction of the distance between the communication device and a nearestbase station of the communication network. For example, thecommunication device may use an initial value of the transmit-targetpower setting parameter 2502A based on a target base station to receivean initial transmission. Thereafter, the communication network mayconfigure the communication device to use a new value of thetransmit-target power setting parameter 2502A based on a serving basestation.

In some embodiments, the transceiver power control loop 2502 may adjustthe transmit-gain setting parameter 2502B in response to changes in biassettings of the transceiver circuitry 2512, a temperature change of thedigital baseband circuit 2510, and/or changes in the voltage standingwave ratio, VSWR, for a particular transmit frequency.

As an example, the transceiver power control loop 2502 may be configuredto adjust a transmit-gain setting parameter 2502B to provide a modulatedradio frequency input signal 2504 to a radio frequency power amplifier2540. In some embodiments, the transmit-gain setting parameter 2502B maybe adjusted as a function of the output power P_(OUT) of the amplifiedradio frequency output signal 2548 provided to the antenna 2544 and thetransmit-target power setting parameter 2502A.

For example, in some embodiments, the transceiver power control loop2502 may increase the transmit-gain setting parameter 2502B used to setthe input power P_(IN) of the modulated radio frequency input signal2504 provided to the radio frequency power amplifier 2540 in response toa decrease in the output power P_(OUT) of an amplified radio frequencyoutput signal 2548 provided to the antenna 2544. In response to theincrease of the transmit-gain setting parameter 2502B, the transceivercircuitry 2512 may increase the drive level of the modulated radiofrequency input signal 2504 provided to the radio frequency poweramplifier 2540. In other words, the transceiver circuitry 2512 mayincrease the input power P_(IN) of the modulated radio frequency inputsignal 2504 to increase the output power P_(OUT) of the amplified radiofrequency output signal 2548. To avoid gain compression due to theincrease drive level of the modulated radio frequency input signal 2504,the envelope tracking control system 2503 may be configured to increasethe amplitude of the modulated power supply voltage, V_(CC), provided tothe collector of the radio frequency power amplifier 2540. Otherwise,the modulated power supply voltage, V_(CC), may be insufficient tosupport the increased output power P_(OUT) to be generated by the radiofrequency power amplifier 2540. Gain compression may result indegradation of the spectrum performance and the error vector magnitude(EVM) performance of the transmitted signal as measured at the output ofthe radio frequency power amplifier 2540.

For example, FIG. 4 depicts a set of iso-gain contours of the radiofrequency power amplifier 2540 that may be stored in a V_(CC) look uptable 2586, and used to generate the modulated power supply voltage,V_(CC), provided to the collector of the radio frequency power amplifier2540. For example, the sets of iso-gain contours may depict an iso-gaincurve for the power gains 28 dBm, 27 dBm, 26 dBm, and 25 dBm. Inaddition, FIG. 4 depicts an original input power P_(IN) or drive levelof a modulated radio frequency input signal provided as an input to theradio frequency power amplifier 2540 to transmit a data signal. FIG. 4further depicts an original range of the modulated power supply voltage,V_(CC), generated by the envelope tracking power converter system 2530as a function of the digital I/Q signal 2506 provided to a digitalbaseband circuit 2510 for transmission. The modulated radio frequencyinput signal 2504 may also be generated based on the digital I/Q signal2506. FIG. 4 further depicts a new range of the modulated power supplyvoltage, V_(CC), that should be generated by the envelope tracking powerconverter system 2530 in response to the input power P_(IN) of amodulated radio frequency input signal being increased by 1 dBm whilemaintaining the same modulation range. As shown in FIG. 4, if theenvelope tracking power converter system 2530 is not configured togenerate a modulated power supply voltage, V_(CC), greater than 3.75volts, the modulated radio frequency input signal may cause the radiofrequency power amplifier 2540 to change from the iso-gain contour for27 dBm of power gain to the iso-gain contour for 26 dBm for an inputpower P_(IN) or drive level above 6 dBm. As a result, the radiofrequency power amplifier 2540 may operate in a non-linear mode and jumpfrom one iso-gain contour to another during the data transmission. Thismay result in gain compression. To avoid this problem, the envelopetracking control system 2503 may determine whether the transceiver powercontrol loop 2502 has increased the drive level of the modulated radiofrequency input signal, such that the envelope tracking power convertersystem 2530 may not be able to provide a sufficient range for themodulated power supply voltage, V_(CC), to support the increase in drivelevel. In that case, the envelope tracking control system 2503 mayconfigure the digital baseband circuit 2510 to generate a control signal2508 that will support the new range of the modulated power supplyvoltage, V_(CC), necessary to prevent gain compression.

As an example, to avoid gain compression, the transceiver power controlloop 2502 may be further configured to cooperatively operate with theenvelope tracking control system 2503. The envelope tracking controlsystem 2503 may provide the control signal 2508 to the envelope trackingpower converter system 2530 as a function of the digital I/Q signal 2506provided to the digital baseband circuit 2510 for transmission. Thedigital I/Q signal 2506 may include a digital in-phase componentI_(DIGITAL) 2506A and a digital quadrature component Q_(DIGITAL) 2506B.In addition, the envelope tracking control system 2503 may be configuredto adjust the control signal 2508 as a function of the transmit-gainsetting parameter 2502B, the transmit-target power setting parameter2502A, a temperature measurement of the digital baseband circuit 2510,and/or some combination thereof. In still other embodiments, theenvelope tracking control system 2503 may be further configured toadjust the control signal 2508 as a function of the transmit frequencyof the transmitter channel for the band of operation in which thecommunication device is configured to operate. As an example, in someembodiments, the envelope tracking control system 2503 may adjust thegeneration of a modulated power supply voltage, V_(CC), as a function ofa transmit-target power setting parameter 2502A and a transmit-gainsetting parameter 2502B, and a transmit frequency of the transmitterchannel.

FIG. 5 may further depict an embodiment of a system, circuitry, andmethods for providing a transceiver power control loop 2502 configuredto regulate the output power P_(OUT) of an amplified radio frequencyoutput signal 2548 provided to the antenna 2544 in response to themodulated radio frequency input signal 2504.

FIG. 5 depicts the envelope tracking modulated power supply system 2500with transceiver power control loop compensation including a digitalbaseband circuit 2510, an envelope tracking power converter system 2530,a radio frequency power amplifier 2540, a radio frequency switch 2542 incombination with an antenna 2544 and power coupler 2546. The digitalbaseband circuit 2510, the envelope tracking power converter system2530, the radio frequency power amplifier 2540, and radio frequencyswitch 2542 may be in communication with a controller 50 via a controlbus 44. The controller 50 may control and/or provide parameters togovern the function of the digital baseband circuit 2510, the envelopetracking power converter system 2530, the radio frequency poweramplifier 2540, and the radio frequency switch 2542 by selectivelysetting parameters and settings in the various devices. In addition, aswill be discussed, the controller 50 may also provide a transmit-targetpower setting parameter 2502A to the digital baseband circuit 2510 asoutput power level control information received by the envelope trackingmodulated power supply system 2500 from a base station of acommunication network in which the envelope tracking modulated powersupply system 2500 is configured to operate. The transmit-target powersetting parameter 2502A may configure the digital baseband circuit 2510to provide a modulated radio frequency input signal 2504 having an inputpower P_(IN) to the radio frequency power amplifier 2540. The digitalbaseband circuit 2510 may be configured to receive the digital I/Qsignal 2506 to be transmitted by the radio frequency power amplifier2540 during a data transmission The digital I/Q signal 2506 may includea digital in-phase component I_(DIGITAL) 2506A and a digital quadraturecomponent Q_(DIGITAL) 2506B. The digital baseband circuit 2510 mayinclude transceiver circuitry 2512 and configuration-feedback circuitry2520. The transceiver circuitry 2512 may include a front-end controlinterface circuit 2514 configured to communicate via a front-end controlinterface 2514A to the envelope tracking power converter system 2530. Inaddition, the digital baseband circuit 2510 may also configure variousoperational parameters of the envelope tracking power converter system2530 via the front-end control interface 2514A. As an example, in someembodiments, the digital baseband circuit 2510 may provide variousoperational parameters via the front-end control interface 2514A to theenvelope tracking power converter system 2530 as a function of aregulated time period.

In addition, the transceiver circuitry 2512 may provide a control signal2508 to the envelope tracking power converter system 2530. In someembodiments, the control signal 2508 may be a differential controlsignal having a non-inverted control signal component 2508A and aninverted control signal component 2508B. For example, the control signal2508 may be a differential V_(RAMP) signal having a non-invertedV_(RAMP) signal component and an inverted V_(RAMP) signal component. Thecontrol signal 2508 may provide a target voltage for the modulated powersupply voltage, V_(CC), at the modulated power supply output 876. Basedon the control signal 2508, the envelope tracking power converter system2530 may generate a modulated power supply voltage, V_(CC), at themodulated power supply output 876. The transceiver circuitry 2512 mayfurther include a temperature sensor 2516 configured to provide atemperature measurement of the digital baseband circuit 2510.

The transceiver circuitry 2512 may also include a feedbackanalog-to-digital converter 2518 configured to receive a measurementsignal 2532 from the envelope tracking power converter system 2530. Themeasurement signal 2532 in combination with the feedbackanalog-to-digital converter 2518 may provide information such as thetemperature and supply voltage of the envelope tracking power convertersystem 2530.

The feedback analog-to-digital converter 2518 may receive themeasurement signal 2532 from the envelope tracking power convertersystem 2530. The feedback analog-to-digital converter 2518 may convertthe measurement signal 2532 into digital data, which may be provided tothe controller 50 or other portions of the digital baseband circuit2510. For example, in some embodiments, the envelope tracking powerconverter system 2530 may provide various types of measurements,including an analog temperature signal, an analog power supply voltagesignal, and a measurement of the battery voltage, V_(BAT).

For example, via the front-end control interface 2514A, the digitalbaseband circuit 2510 may request the envelope tracking power convertersystem 2530 to provide a measurement of the battery voltage, V_(BAT),from a battery 20. In response, the envelope tracking power convertersystem 2530 may provide a measurement of the battery voltage, V_(BAT),to the feedback analog-to-digital converter 2518 via the measurementsignal 2532. In a similar fashion, the digital baseband circuit 2510 mayobtain a measurement of the temperature of the envelope tracking powerconverter system 2530 via the measurement signal 2532.

Based on the digital in-phase component I_(DIGITAL) 2506A and thedigital quadrature component Q_(DIGITAL) 2506B, the transceivercircuitry 2512 may generate a control signal 2508 to provide a targetvoltage for generation of a modulated power supply voltage, V_(CC). Aspreviously described, in some embodiments the control signal 2508 may bea differential V_(RAMP) signal. The envelope tracking power convertersystem 2530 may configure the digital baseband circuit 2510 to generatethe control signal 2508 as a function of the modulated radio frequencyinput signal 2504 provided to the radio frequency power amplifier 2540for transmission. The envelope tracking power converter system 2530 maygenerate the modulated power supply voltage, V_(CC), at the modulatedpower supply output 876 in response to the control signal 2508. Theenvelope tracking power converter system 2530 may provide the modulatedpower supply voltage, V_(CC), to the radio frequency power amplifier2540.

The radio frequency power amplifier 2540 may generate an amplified radiofrequency output signal 2548 as a function of the input power P_(IN) ofthe modulated radio frequency input signal 2504 and the modulated powersupply voltage V_(CC). The radio frequency power amplifier 2540 mayprovide the amplified radio frequency output signal 2548 to the antenna2544 via a radio frequency switch 2542.

In response to the amplified radio frequency output signal 2548, thepower coupler 2546 may obtain a measurement of the amplified radiofrequency output signal 2548. In some embodiments, the power coupler2546 may also obtain a measurement of the output power of the amplifiedradio frequency output signal 2548, present at the antenna 2544. Inother embodiments of the transceiver power control loop 2502, the powercoupler 2546 may be located at the output of the radio frequency poweramplifier 2540 and configured to obtain a measurement of the amplifiedradio frequency output signal 2548 before passing through the radiofrequency switch 2542. Some embodiments of the power coupler 2546 mayinclude a power detector configured to measure the output power P_(OUT)of the amplified radio frequency output signal 2548 at the output of theradio frequency power amplifier 2540. The power coupler 2546 may providea measured amplified radio frequency output signal 2550 to the digitalbaseband circuit 2510. For example, the power coupler 2546 may providethe measured amplified radio frequency output signal 2550 to theconfiguration-feedback circuitry 2520 of the digital baseband circuit2510.

The I/Q feedback system 2522 may receive the measured amplified radiofrequency output signal 2550. The I/Q feedback system 2522 maydemodulate the measured amplified radio frequency output signal 2550 toobtain a digitized in-phase feedback component I_(MEASURED) 2570 and adigitized quadrature feedback component Q_(MEASURED) 2572 as a functionof the measured amplified radio frequency output signal 2550.

The digitized in-phase feedback component I_(MEASURED) 2570 and thedigitized quadrature feedback component Q_(MEASURED) 2572 may beprovided to the fast calibration iso-gain and delay alignment circuitry2524 and the transceiver power control loop circuitry 2526. Thetransceiver power control loop circuitry 2526 also receives thetransmit-target power setting parameter 2502A. Based on the digitizedin-phase feedback component I_(MEASURED) 2570 and the digitizedquadrature feedback component Q_(MEASURED) 2572, the transceiver powercontrol loop circuitry 2526 may determine whether to increase ordecrease the power level of the modulated radio frequency input signal2504 as a function of the transmit-target power setting parameter 2502A.

For example, the transceiver power control loop circuitry 2526 maydetermine that the output power P_(OUT) of the measured amplified radiofrequency output signal 2550 is less than the transmit-target powersetting parameter 2502A. In response to the determination that theoutput power P_(OUT) of the measured amplified radio frequency outputsignal 2550 is less than the transmit-target power setting parameter2502A, the transceiver power control loop circuitry 2526 may configurethe transceiver circuitry 2512 to increase the input power P_(IN) of themodulated radio frequency input signal 2504 provided to the radiofrequency power amplifier 2540. For example, in some embodiments, thetransceiver power control loop circuitry 2526 may increase the magnitudeof the transmit-gain setting parameter 2502B. The transceiver powercontrol loop circuitry 2526 may generate a transmit-gain setting signal2526A based on the transmit-gain setting parameter 2502B. In someembodiments, the transceiver power control loop circuitry 2526 mayprovide the transmit-gain setting signal 2526A to a baseband controller2511. The baseband controller 2511 may be configured to adjust variousparameters and settings of the transceiver circuitry 2512 to increasethe drive level or input power P_(IN) of the modulated radio frequencyinput signal 2504 to compensate for the difference between thetransmit-target power setting parameter 2502A and the output powerP_(OUT) of the measured amplified radio frequency output signal 2550.

In addition, the transceiver power control loop circuitry 2526 mayprovide the transmit-gain setting signal 2526A and a transmit-targetpower setting signal 2526B to a pseudo-envelope follower gain controlcircuit 2574. Alternatively, in some embodiments, the basebandcontroller 2511 may provide the transmit-gain setting parameter 2502Band a transmit-target output power parameter to the pseudo-envelopefollower gain control circuit 2574. For example, in some alternativeembodiments, the baseband controller 2511 may configure thepseudo-envelope follower gain control circuit 2574 to generate a gaincontrol signal 2574A having a gain control magnitude substantially equalto G_(CONTROL).

On the other hand, the transceiver power control loop circuitry 2526 maydetermine that the output power P_(OUT) of the measured amplified radiofrequency output signal 2550 exceeds the transmit-target power settingparameter 2502A. Based on a determination that the output power P_(OUT)of the measured amplified radio frequency output signal 2550 exceeds thetransmit-target power setting parameter 2502A, the transceiver powercontrol loop circuitry 2526 may decrease the magnitude of thetransmit-gain setting parameter 2502B.

The transceiver power control loop circuitry 2526 may generate atransmit-gain setting signal 2526A based on the reduced magnitude of thetransmit-gain setting parameter 2502B. In some embodiments, the basebandcontroller 2511 may be configured to adjust various parameters andsettings of the transceiver circuitry 2512 to reduce the drive level orinput power P_(IN) of the modulated radio frequency input signal 2504 todecrease the output power P_(OUT) of the measured amplified radiofrequency output signal 2550. In addition, the transceiver power controlloop circuitry 2526 may provide the transmit-gain setting signal 2526Ato the pseudo-envelope follower gain control circuit 2574.

In some embodiments the power coupler 2546 may include a detector. Thedetector may include a sensor in communication with an output of theradio frequency power amplifier 2540. The detector may measure an outputpower generated by the radio frequency power amplifier 2540. As anexample, after accounting for losses in the radio frequency switch 2542,the power coupler 2546 may provide a measured output power, P_(OUT), ofthe amplified radio frequency output signal 2548. The measured outputpower may provide an average power measurement. The power coupler 2546may also provide an average power measurement based on various averagingtechniques. In some embodiments, the power coupler 2546 may beconfigured to provide a peak power measurement. In some embodiments, thepower coupler 2546 may be configured as a linear power detector. Inother embodiments, the power coupler 2546 may be a logarithmic powerdetector. In still other embodiments, the power coupler 2546 may be anon-linear power detector.

The radio frequency switch 2542 includes a closed state of operation andan open state of operation. In the closed state of operation, the radiofrequency switch 2542 may couple the radio frequency power amplifier2540 to the antenna 2544, which permits the amplified radio frequencyoutput signal 2548 to pass through the radio frequency switch 2542 tothe antenna 2544 to transmit the amplified radio frequency output signal2548. In the open state of operation, the radio frequency switch 2542substantially attenuates the amplified radio frequency output signal2548 to minimize or prevent a transmission of the amplified radiofrequency output signal 2548. Ideally, in the open state of operation,the radio frequency switch 2542 has infinite impedance. In someembodiments, the radio frequency switch 2542 may provide antenna switchisolation around +25 dBm.

In some embodiments, the communication device may include a controller50 configured to coordinate the functions of the various components andoperations of the embodiments of the envelope tracking modulated powersupply system 2500 with transceiver power control loop compensation. Insome embodiments, the controller 50 may include a number of distributedprocessors, distributed microcontrollers, controllers, local controlunits, memory mapped local memory for various components, functions, andfeatures of the various embodiments of the power management system 870,the various embodiments of the switch mode power supply converter 872,the various embodiments of the parallel amplifier circuit 874, firmwarecircuitry, reconfigurable digital logic circuitry, state machines,analog logic circuitry, a number of communication interfaces and busses,various forms of memory, data registers, control registers, cachememory, distributed memory, memory mappings, register mappings, varioustypes of sensors and inputs for receiving sensory data or information,one or more digital-to-analog converters, one or more analog-to-digitalconverters, various types and numbers of output drivers, various typesand numbers of digital input buffers and analog input buffers, variousprocessor cores and arithmetic operation units, sub-processors, readonly memory, random access memory, flash memory circuitry,electronically fusible memory, interrupt handlers, interrupt handlingmanagement systems, and controls signals. For the sake of simplicity ofdescription, and not by way of limitation, the controller 50 may bedescribed as performing various functions and features to govern theoperation of the communication device, the various embodiments of theenvelope tracking modulated power supply system 2500 with transceiverpower control loop compensation, the digital baseband circuit 2510, theenvelope tracking power converter system 2530, the radio frequency poweramplifier 2540, the radio frequency switch 2542, the power coupler 2546.As an example, and not by way of limitation, the controller 50 mayinclude a control bus 44 in communication with the digital basebandcircuit 2510, the envelope tracking power converter system 2530, theradio frequency power amplifier 2540, the radio frequency switch 2542,and the power coupler 2546. For the sake of simplicity, and not by wayof limitation, the control bus 44 is only depicted as being incommunication with each component depicted in FIG. 1. Even so, one ofordinary skill in the art will understand that this is by way of exampleand not by way of limitation.

For example, the controller 50 may be in communication with variousother elements to be described in the digital baseband circuit 2510, theembodiments of the switch mode power supply converter 872, and theembodiments of the parallel amplifier circuit 874 of the envelopetracking power converter system 2530. As an example, the controller 50may be in communication with and cooperatively operate with the variousembodiments of the power management system 870.

The controller 50 may include a processor, a Digital Signal Processor(DSP), an Application Specific Integrated Circuit (ASIC), a FieldProgrammable Gate Array (FPGA) or other programmable logic device,discrete gate or transistor logic, discrete hardware components, or anycombination thereof designed to perform the functions described herein.A processor may be a microprocessor. In the alternative, the processormay be any conventional processor, controller, microcontroller, a statemachine and/or a combination thereof. A processor may also beimplemented as a combination of computing devices. As an example, acombination of computing devices may include a combination of a DSP anda microprocessor, a plurality of microprocessors, one or moremicroprocessors in conjunction with a DSP core, or any other suchconfiguration. The processor 28 may further include or be embodied inhardware and in computer executable instructions that are stored inmemory, and may reside, for example, in Random Access Memory (RAM),flash memory, Read Only Memory (ROM), Electrically Programmable ROM(EPROM), Electrically Erasable Programmable ROM (EEPROM), registers,hard disk, a removable disk, a CD-ROM, or any other form of computerreadable medium known in the art. An exemplary storage medium may becoupled to the processor such that a processor can read informationfrom, and write information to, the storage medium. In the alternative,the storage medium or a portion of the storage medium may be integral tothe processor. The processor and the storage medium may reside in anApplication Specific Integrated Circuit (ASIC).

The front-end control interface circuit 2514 is in communication withthe envelope tracking power converter system 2530 via a front-endcontrol interface 2536. In some embodiments, the front-end controlinterface circuit 2514 may operate with a Mobile Industry ProcessorInterface® standard based device via the front-end control interface2536. As an example, the front-end control interface circuit 2514 andthe front-end control interface 2536 may be configured to controldevices based on a Radio Frequency Front-End control interfacespecification.

In some embodiments, the controller 50 may cooperatively interoperatethe front-end control interface circuit 2514 to configure and controlthe envelope tracking power converter system 2530. In still otherembodiments, the controller 50 may be a supervisory processor thatgoverns other controllers. The baseband controller 2511 may includeconfigurable firmware, a processor, sub-controllers, control logic,control circuitry, one or more state machines, a Digital SignalProcessor (DSP), an ASIC, a Field Programmable Gate Array (FPGA) orother programmable logic device, discrete gate or transistor logic,discrete hardware components, or any combination thereof designed toperform the functions described herein. The baseband controller 2511 mayinclude firmware, executable program code, read only memory, fusiblememory, a memory processor, a microprocessor, conventional processor,controller, microcontroller, or state machine. A processor may also beimplemented as a combination of computing devices. As an example, acombination of computing devices may include a combination of a DSP anda microprocessor, a plurality of microprocessors, one or moremicroprocessors in conjunction with a DSP core, or any other suchconfiguration. The controller 50 may further include or be embodied inhardware and in computer executable instructions that are stored inmemory, and may reside, for example, in Random Access Memory (RAM),flash memory, Read Only Memory (ROM), Electrically Programmable ROM(EPROM), Electrically Erasable Programmable ROM (EEPROM), registers,hard disk, a removable disk, a CD-ROM, or any other form of computerreadable medium known in the art. An exemplary storage medium may becoupled to the processor such that a processor can read informationfrom, and write information to, the storage medium. In the alternative,the storage medium or a portion of the storage medium may be integral tothe processor. The processor and the storage medium may reside in anApplication Specific Integrated Circuit (ASIC).

The digital baseband circuit 2510 may receive a digital I/Q signal 2506to be transmitted by the radio frequency power amplifier 2540. Forexample, the digital I/Q signal 2506 includes a digital data signal.Alternatively, the digital I/Q signal 2506 may be a test signal forcalibrating various parts of the envelope tracking modulated powersupply system 2500 with transceiver power control loop compensation. Insome embodiments, the digital I/Q signal 2506 may include a digitalin-phase component I_(DIGITAL) 2506A and a digital quadrature componentQ_(DIGITAL) 2506B. The digital in-phase component I_(DIGITAL) 2506A andthe digital quadrature component Q_(DIGITAL) 2506B where the digitalin-phase component I_(DIGITAL) 2506A and the digital quadraturecomponent Q_(DIGITAL) 2506B represent the in-phase and quadraturecomponents of a signal to be transmitted. The digital I/Q signal 2506may receive various types of test signals to generate a modulated radiofrequency input signal 2504 having a desired envelope characteristic. Inaddition, the digital I/Q signal 2506 may include a test signal thatgenerates the modulated radio frequency input signal 2504 having adesired frequency characteristic or periodicity. For example, dependentupon a band of operation of a communication network to which acommunication device is assigned or a wideband modulation techniqueassociated with the band of operation, the test signal received by thedigital I/Q signal 2506 may include various frequency characteristics.As an example, for the case where a wideband modulation techniqueincludes a particular modulation bandwidth, the digital I/Q signal 2506may generate a modulated radio frequency input signal 2504 that has abandwidth substantially similar to the bandwidth associated with thewideband modulation technique or the band of operation assigned to thecommunication device.

The digital baseband circuit 2510 may further include a pseudo-envelopefollower gain control circuit 2574 configured to provide a gain controlsignal 2574A having a gain control magnitude substantially equal toG_(CONTROL). The pseudo-envelope follower gain control circuit 2574 mayadjust the magnitude of the gain control signal 2574A based as afunction of a measured temperature of the transceiver circuitry 2512,the transmit-gain setting signal 2526A, a carrier frequency for the bandof operation in which the communication device is configured to operate,and/or a combination thereof to generate the V_(CC) look up table indexsignal 2584 at the input of the V_(CC) look up table 2586. Thepseudo-envelope follower gain control circuit 2574 may adjust the gaincontrol magnitude, G_(CONTROL), such that the V_(CC) look up table indexsignal 2584 provides an index signal into the V_(CC) look up table 2586that substantially emulates the drive level or input power P_(IN) of themodulated radio frequency input signal 2504 provided to the radiofrequency power amplifier 2540. In other words, the pseudo-envelopefollower gain control circuit 2574 may adjust the magnitude of the gaincontrol magnitude, G_(CONTROL), such that the V_(CC) look up tablecompensated envelope signal 2588 may provide a target voltage for themodulated power supply voltage, V_(CC), as a function of the iso-gaincurve that corresponds to the transmit-target power setting parameter2502A. As a result, the gain control signal 2574A may compensate for anincrease in the input power of the modulated radio frequency inputsignal provided to the radio frequency power amplifier 2540 to preventor minimize gain compression.

The envelope tracking power converter system 2530 may include variouscircuits to generate a modulated power supply voltage, V_(CC), at themodulated power supply output 876. As a non-limiting example, someembodiments of the envelope tracking power converter system 2530 mayinclude the switch mode power supply converter 872 and the parallelamplifier circuit 874. In some embodiments, the switch mode power supplyconverter 872 may be a buck converter. In other embodiments, the switchmode power supply converter 872 may include a multi-level charge pumpbuck converter configured to buck and/or boost a supply voltage toprovide a switching voltage at different voltages dependent upon thecontrol signal 2508. As a non-limiting example, in some embodiments, theenvelope tracking power converter system 2530 may operate in variouspower level modulation modes depending on the expected output power tobe generated by the radio frequency power amplifier 2540 in a datatransmission.

As depicted in FIG. 6, the transceiver circuitry 2512 may include anAM/PM compensation circuit 2573, a digital gain control circuit 2576, anenvelope magnitude calculation circuit 2578, and an envelope trackingdigital/analog section 2580. The envelope magnitude calculation circuit2578 may receive the digital in-phase component I_(DIGITAL) 2506A andthe digital quadrature component Q_(DIGITAL) 2506B. The envelopemagnitude calculation circuit 2578 may calculate an envelope magnitude,ζ_(MAGNITUDE), of an expected envelope of the modulated radio frequencyinput signal 2504, P_(IN). As an example, the envelope magnitudecalculation circuit 2578 may generate the envelope magnitude,ζ_(MAGNITUDE), based on the digital in-phase component I_(DIGITAL) 2506Aand the digital quadrature component Q_(DIGITAL) 2506B. In someembodiments, the envelope magnitude, ζ_(MAGNITUDE), may be proportionalto the magnitude of the envelope of the modulated radio frequency inputsignal 2504, P_(IN). As a non-limiting example, the envelope magnitude,ζ_(MAGNITUDE), may be described by equation (1) as follows:ζ_(MAGNITUDE)=√{square root over (I _(DIGITAL) ² +Q _(DIGITAL) ²)}  (1)The envelope magnitude calculation circuit 2578 provides the envelopemagnitude, ζ_(MAGNITUDE), to the envelope tracking digital/analogsection 2580.

The AM/PM compensation circuit 2573 may be configured to receive thedigital in-phase component I_(DIGITAL) 2506A and the digital quadraturecomponent Q_(DIGITAL) 2506B. In some embodiments, the AM/PM compensationcircuit 2573 may be configured to provide amplitude modulation and phasemodulation compensation for the radio frequency power amplifier 2540.For example, some embodiments of the AM/PM compensation circuit 2573 mayadjust the digital in-phase component I_(DIGITAL) 2506A and the digitalquadrature component Q_(DIGITAL) 2506B to compensate for any phase shiftdue to the modulated power supply voltage, V_(CC), the input power,P_(IN), of the modulated radio frequency input signal 2504 to betransmitted by the radio frequency power amplifier 2540, and/or acombination thereof.

For example, some embodiments of the AM/PM compensation circuit 2573 mayreceive the V_(CC) look up table index signal 2584 and the V_(CC) lookup table compensated envelope signal 2588. The V_(CC) look up tableindex signal 2584 may be substantially equivalent to a digitalrepresentation of the envelope of the modulated radio frequency inputsignal 2504 provided to the radio frequency power amplifier 2540 fortransmission. The V_(CC) look up table compensated envelope signal 2588may be substantially equivalent to a digital representation of a targetenvelope for the envelope tracking power converter system 2530 to trackor follow to generate a modulated power supply voltage, V_(CC), thatsubstantially corresponds to the modulated radio frequency input signal2504. In other words, the V_(CC) look up table 2586 may translate theV_(CC) look up table index signal 2584 into a desired target voltage forthe modulated power supply voltage, V_(CC).

As will be described, the pseudo-envelope follower gain control circuit2574 may generate the gain control, G_(CONTROL), signal as a function ofa temperature of the digital baseband circuit 2510, the transmit-targetpower setting signal 2526B, the transmit-gain setting signal 2526A,and/or a combination thereof to adjust the magnitude of the envelopemagnitude, ζ_(MAGNITUDE), signal. As a result, the V_(CC) look up tableindex signal 2584 may include a transceiver gain adjusted envelopemagnitude ζ_(TGA-MAGNITUDE), substantially equal to the product of theenvelope magnitude, ζ_(MAGNITUDE), signal and the gain control,G_(CONTROL), signal. The gain control, G_(CONTROL), adjusts the envelopemagnitude, ζ_(MAGNITUDE), signal to reflect a change in the transceivergain such that the V_(CC) look up table index signal is not solely basedon the expected output power to be generated by the radio frequencypower amplifier 2540 in response to the modulated radio frequency inputsignal 2504. Transceiver gain adjusted envelope magnitude,ζ_(TGA-MAGNITUDE), is provided by the equation (2) as follows:ζ_(TGA-MAGNITUDE)=ζ_(MAGNITUDE) ×G _(CONTROL),  (2)where the gain control, G_(CONTROL), may be a function of thetransceiver temperature, T_(TRANSCEIVER), and the transceiver gainsettings, G_(TRANSCEIVER). In some embodiments, the gain control,G_(CONTROL), may also be a function of the transceiver transmitfrequency, F_(TRANSMIT).

Accordingly, the V_(CC) look up table index signal 2584 substantiallycompensates for a change in the gain of the digital transceiver toreflect the input drive level to the radio frequency power amplifier.Thus, the V_(CC) look up table index signal 2584 provides an emulated“Pin drive level of the power amplifier” rather using as index theoutput power of the radio frequency power amplifier 2540 to generate thecontrol signal 2508.

Because the gain control, G_(CONTROL), provided by the pseudo-envelopefollower gain control circuit 2574 compensates for changes in thetransmit gain setting used to set the power level of the modulated radiofrequency input signal 2504, the V_(CC) look up table index signal 2584may index into the V_(CC) look up table 2586 to generate a V_(CC) lookup table compensated envelope signal 2588 that is adjusted for a changein the transmit-gain setting signal 2526A used to set the power level ofthe modulated radio frequency input signal 2504 provided to the radiofrequency power amplifier 2540.

The AM/PM compensation circuit 2573 may compensate for any phase shiftdue to the drive level change in of the modulated radio frequency inputsignal 2504 provided to the radio frequency power amplifier 2540.

In some embodiments, the AM/PM compensation circuit 2573 may receiveeither the V_(CC) look up table index signal 2584 or the V_(CC) look uptable compensated envelope signal 2588 as a scaled version of thedigital envelope of the modulated radio frequency input signal 2504. Inother words, the AM/PM compensation circuit 2573 may use either theV_(CC) look up table index signal 2584, the V_(CC) look up tablecompensated envelope signal 2588, and/or a combination thereof torepresent the modulated radio frequency input signal 2504. Based on therepresentation of the modulated radio frequency input signal 2504, theAM/PM compensation circuit 2573 may calculate a phase correction as afunction of the input power P_(IN). The AM/PM compensation circuit 2573may apply the phase correction to generate a phase corrected digitalin-phase component I_(DIGITAL) 2573I and a phase corrected digitalin-phase component I-DIGITAL 2527Q, which are provide to the digitalgain control circuit 2576. In response to the phase corrected digitalin-phase component I_(DIGITAL) 2573I and the phase corrected digitalin-phase component I_(DIGITAL) 2527Q, the digital gain control circuit2576 may generate a gain adjusted digital in-phase, I_(DIGITAL) _(—)_(GAIN) _(—) _(ADJUSTED), component and the gain adjusted digitalquadrature, Q_(DIGITAL) _(—) _(GAIN) _(—) _(ADJUSTED), component basedon the transmit-gain setting parameter 2502B.

The envelope tracking digital/analog section 2580 may include thepseudo-envelope follower gain control circuit 2574, a multiplier 2582,the V_(CC) look up table 2586, a fine tuning delay circuit 2590, adigital pre-distortion compensation filter 2594, a differentialdigital-to-analog converter 2598, and a fixed delay circuit 2600. Themultiplier 2582 may include a first input in communication with theoutput of the pseudo-envelope follower gain control circuit 2574 and asecond input in communication with the envelope magnitude calculationcircuit 2578. The first input of the multiplier 2582 may receive thegain control, G_(CONTROL), from the pseudo-envelope follower gaincontrol circuit 2574. The second input of the multiplier 2582 mayreceive the envelope magnitude, ζ_(MAGNITUDE). The multiplier 2582multiplies the envelope magnitude, ζ—_(MAGNITUDE), by the gain control,G_(CONTROL), to generate the V_(CC) look up table index signal 2584. Insome embodiments, the gain control, G_(CONTROL), is adjusted tocompensate for variations in a setup of the digital baseband circuit2510. For example, the value of the gain control, G_(CONTROL), may beadjusted to increase or decrease the modulated power supply voltage,V_(CC), generated by the envelope tracking power converter system 2530as a function of the input power P_(IN) or drive level of the modulatedradio frequency input signal 2504 provided to the radio frequency poweramplifier 2540.

The multiplier 2582 provides a V_(CC) look up table index signal 2584 tothe V_(CC) look up table 2586 based on the product of the envelopemagnitude, ζ_(MAGNITUDE), and the gain control, G_(CONTROL).

The V_(CC) look up table 2586 may include one or more tables totranslate the V_(CC) look up table index signal 2584 into a V_(CC) lookup table compensated envelope signal 2588. The V_(CC) look up tablecompensated envelope signal 2588 may provide a target voltage for themodulated power supply voltage, V_(CC), to be generated by the envelopetracking power converter system 2530 based on the envelope of themodulated radio frequency input signal 2504. For example, depending upona band of operation, a power level modulation mode of operation of thecommunication device, a spectrum target, and/or an error vectormagnitude target, the baseband controller 2511 or the controller 50 mayconfigure the V_(CC) look up table 2586 to select a particular set oflook up values as a function of an operational mode of the communicationdevice, the digital baseband circuit 2510, the envelope tracking powerconverter system 2530, the envelope tracking digital/analog section2580, and/or a combination thereof. The V_(CC) look up table 2586provides a V_(CC) look up table compensated envelope signal 2588 to thefine tuning delay circuit 2590.

The fine tuning delay circuit 2590 may be configured to delaypropagation of the V_(CC) look up table compensated envelope signal 2588based on a programmable delay parameter. The programmable delayparameter may be controlled by the controller 50 the baseband controller2511 and/or a combination thereof. Depending upon the programmable delayparameter, the fine tuning delay circuit 2590 may provide a delaypropagation of the V_(CC) look up table compensated envelope signal 2588based on a programmable delay parameter by a fine tuning delay time,T_(FINE) _(—) _(TUNING). The granularity of the fine tuning delay time,T_(FINE) _(—) _(TUNING), may depend upon a clock of the digital basebandcircuit 2510. In some embodiments, the propagation delay of the finetuning delay circuit 2590 may be provided by an interpolator. In otherembodiments, the propagation delay of the fine tuning delay circuit 2590may be provided by a tapped delay line or other delay circuit. The finetuning delay circuit 2590 provides a digitally delayed control signalwithout pre-distortion compensation 84 to the digital pre-distortioncompensation filter 2594. The digitally delayed control signal withoutpre-distortion compensation 84 may be provided to the digitalpre-distortion compensation filter 2594.

The digital pre-distortion compensation filter 2594 may be either aninfinite impulse response filter (IIR) or a finite impulse responsefilter (FIR). In some embodiments, the coefficients of the digitalpre-distortion compensation filter 2594 may be configured by thecontroller 50, baseband controller 2511 and/or a combination thereof. Inaddition, the propagation delay through the digital pre-distortioncompensation filter 2594 may vary depending upon the various modes ofoperation of the communication device.

For example, the propagation delay through the digital pre-distortioncompensation filter 2594 may depend on various factors including a bandof operation, a channel bandwidth, a wideband modulation bandwidth,channel conditions, a location of the communication device relative tocommunication hubs, points, towers, and hot-spots, signal modulationtechniques, expected envelope characteristics, the peak-to-average ratioof the envelope of the signal to be transmitted, the data rate, thebandwidth of the channel, error vector magnitude, desired transmitspectrum characteristics, and/or some combination thereof.

As an example application, the digital pre-distortion compensationfilter 2594 may be configured to equalize the frequency response of themodulated power supply voltage, V_(CC), relative to the differentialV_(RAMP) signal. The digital pre-distortion compensation filter 2594 mayprovide a digital control signal 2596 to the differentialdigital-to-analog converter 2598. In some embodiments, the digitalcontrol signal 2596 may be a digital V_(RAMP) signal, V_(RAMP) _(—)_(DIGITAL). The differential digital-to-analog converter 2598 mayconvert the digital control signal 2596 into the control signal 2508. Insome embodiments, the control signal 2508 may be a differential controlsignal. For example, as described above, the control signal 2508 mayinclude the non-inverted control signal component 2508A and the invertedcontrol signal component 2508B. In some embodiments, the non-invertedcontrol signal component 2508A may be the non-inverted V_(RAMP) signalcomponent, V_(RAMP)+, and the inverted control signal component 2508Bmay be the inverted V_(RAMP) signal component, V_(RAMP)−. Although notdepicted in FIG. 6, the differential digital-to-analog converter 2598may further include anti-aliasing filters to accomplish proper imagerejection after converting digital data into an analog signal.

The fixed delay circuit 2600 may provide a fixed delay to offset thepropagation delay of the of the envelope magnitude, ζ_(MAGNITUDE),signal through the envelope tracking digital/analog section 2580 andprovide a range of delay for the fine tuning delay circuit 2590. Thefixed delay circuit 2600 receives the gain adjusted digital in-phase,I_(DIGITAL) _(—) _(GAIN) _(—) _(ADJUSTED), component and the gainadjusted digital quadrature, Q_(DIGITAL) _(—) _(GAIN) _(—) _(ADJUSTED),component from the digital gain control circuit 2576. The fixed delaycircuit 2600 provides a delay adjusted digital in-phase component 2604and a delay adjusted quadrature component 2618 as outputs.

A first digital-to-analog converter circuit 2606 is configured toreceive the delay adjusted digital in-phase component 2604. The firstdigital-to-analog converter circuit 2606 converts the delay adjusteddigital in-phase component 2604 into an analog signal to generate apre-reconstruction in-phase component 2608, which is provided to a firstanti-aliasing filter 2610. The first anti-aliasing filter 2610 filtersthe pre-reconstruction in-phase component 2608 to provide an analogin-phase component 2612 to be mixed.

Similarly, a second digital-to-analog converter circuit 2620 may receivethe delay adjusted quadrature component 2618. The seconddigital-to-analog converter circuit 2620 converts the delay adjustedquadrature component 2618 into a pre-reconstruction quadrature component2622, which is provided to a second anti-aliasing filter 2624. Thesecond anti-aliasing filter 2624 filters the pre-reconstructionquadrature component 2622 to provide an analog quadrature component 2626to be mixed.

The digital baseband circuit 2510 may further include an oscillator2558, an in-phase component mixer 2614, I-Mixer, and a quadraturecomponent mixer 2628, Q-Mixer. The oscillator 2558 is configured togenerate a carrier signal 2558A having a transmit carrier frequency fora band of operation, which is provided to the quadrature component mixer2628, Q-Mixer, and the in-phase component mixer 2614, I-Mixer. Thequadrature component mixer 2628, Q-Mixer, mixes the analog quadraturecomponent 2626 to generate a modulated quadrature component 2630. Thein-phase component mixer 2614, I-Mixer, mixes the analog in-phasecomponent 2612 to generate a modulated in-phase component 2616. Themodulated quadrature component 2630 and the modulated in-phase component2616 are provided to a radio frequency attenuator 2632. The controller50 may adjust the radio frequency attenuator 2632 to provide a magnitudeof the modulated quadrature component 2630 and the modulated in-phasecomponent 2616 used to provide the modulated radio frequency inputsignal 2504 having an input power P_(IN) to the radio frequency poweramplifier 2540. The radio frequency attenuator 2632 may be either adigital radio frequency attenuator or an analog radio frequencyattenuator.

The fine tuning delay time, T_(FINE) _(—) _(TUNING), of the fine tuningdelay circuit 2590 may be configured to adjust the group delayrelationship between the control signal 2508 provided to the envelopetracking power converter system 2530 and the modulated radio frequencyinput signal 2504, P_(IN), provided to the radio frequency poweramplifier 2540. The output power of the amplified radio frequency outputsignal 2548 may depend on the relative group delay mismatch between thecontrol signal 2508 and the modulated radio frequency input signal 2504,P_(IN).

FIG. 6 depicts a more detailed view of the digital baseband circuit 2510including the baseband controller 2511, the transceiver circuitry 2512,and the configuration-feedback circuitry 2520. Although FIG. 6 depictsthat the transceiver circuitry 2512 and the configuration-feedbackcircuitry 2520 are substantially separate portions of the digitalbaseband circuit 2510, in some embodiments, the features and thefunctions of the transceiver circuitry 2512 and theconfiguration-feedback circuitry 2520 may be cooperatively shared.

FIG. 6 further depicts an I/Q feedback system 2522 having a buffer 2552configured to receive the measured amplified radio frequency outputsignal 2550 from the power coupler 2546. The buffer 2552 provides themeasured amplified radio frequency output signal 2550 to the in-phasecomponent feedback mixer 2554 and the quadrature component feedbackmixer 2556. The in-phase component feedback mixer 2554 and thequadrature component feedback mixer 2556 may receive a carrier signal2558A having a transmit carrier frequency from an oscillator 2558. Thein-phase component feedback mixer 2554 and the quadrature componentfeedback mixer 2556 may demodulate the measured amplified radiofrequency output signal 2550 to recover an in-phase feedback component2562 and a quadrature feedback component 2564. The in-phase componentfeedback mixer 2554 may provide the in-phase feedback component 2562 toan in-phase component analog-to-digital converter 2566 to generate adigitized in-phase feedback component I_(MEASURED) 2570. The quadraturecomponent feedback mixer 2556 may provide the quadrature feedbackcomponent 2564 to a quadrature component analog-to-digital converter2568 to generate a digitized quadrature feedback component Q_(MEASURED)2572. The fast calibration iso-gain and delay alignment circuitry 2524and the transceiver power control loop circuitry 2526 may each receivethe digitized in-phase feedback component I_(MEASURED) 2570 and thedigitized quadrature feedback component Q_(MEASURED) 2572.

As a non-limiting example, the V_(CC) look up table 2586 may store aplurality of calibrated iso-gain contours. The iso-gain contours may becalibrated by measuring the relationship between a range of drive levelsor input power P_(IN) levels of a modulated radio frequency input signal2504 versus an output power P_(OUT) generated by the radio frequencypower amplifier 2540 at a given modulated power supply voltage, V_(CC).For example, in the simplest form, the input power P_(IN) for amodulated radio frequency input signal 2504 may sweep through 27 dBn to+7 dbm at a given modulated power supply voltage, Vcc. The fastcalibration iso-gain and delay alignment circuitry 2524 may determinethe output power generated in response to the input power P_(IN) level.The process may then be repeated for different magnitudes of themodulated power supply voltage, V_(CC).

One embodiment of the envelope tracking modulated power supply system2500 as illustrated in FIGS. 5 and 6 is presented. Theconfiguration-feedback circuitry 2520 regulates an output power from theradio frequency power amplifier 2540 based on a difference between atarget output power from the radio frequency power amplifier 2540 and ameasured output power from the radio frequency power amplifier 2540. Thetransceiver circuitry 2512 regulates the modulated power supply voltage,V_(CC), which is used by the radio frequency power amplifier 2540 toprovide power for amplification, based on the difference between thetarget output power from the radio frequency power amplifier 2540 andthe measured output power from the radio frequency power amplifier 2540.

In one embodiment of the envelope tracking modulated power supply system2500, the configuration-feedback circuitry 2520 regulates the outputpower from the radio frequency power amplifier 2540, such that themeasured output power from the radio frequency power amplifier 2540 isabout equal to the target output power from the radio frequency poweramplifier 2540. The transceiver circuitry 2512 provides the modulatedradio frequency input signal 2504 having the input power P_(IN) to theradio frequency power amplifier 2540. The radio frequency poweramplifier 2540 generates the amplified radio frequency output signal2548 as a function of the input power P_(IN) of the modulated radiofrequency input signal 2504 and the modulated power supply voltageV_(CC). The measured output power from the radio frequency poweramplifier 2540 is representative of the output power from the radiofrequency power amplifier 2540. The measured amplified radio frequencyoutput signal 2550 is representative of the measured output power fromthe radio frequency power amplifier 2540.

In one embodiment of the envelope tracking modulated power supply system2500, the transmit-gain setting signal 2526A is based on thetransmit-gain setting parameter 2502B and the transmit-target powersetting signal 2526B is based on the transmit-target power settingparameter 2502A. As such, the transmit-target power setting parameter2502A is representative of the target output power from the radiofrequency power amplifier 2540 and the transmit-gain setting parameter2502B is representative of the transmit-gain setting of the transceivercircuitry 2512. In this regard, the configuration-feedback circuitry2520 receives the measured amplified radio frequency output signal 2550and provides the transmit-gain setting parameter 2502B based on themeasured amplified radio frequency output signal 2550 and thetransmit-target power setting parameter 2502A. As a result, thetransmit-gain setting signal 2526A is based on the measured amplifiedradio frequency output signal 2550 and the transmit-target power settingparameter 2502A.

In one embodiment of the envelope tracking modulated power supply system2500, the transceiver circuitry 2512 adjusts the input power P_(IN) ofthe modulated radio frequency input signal 2504 provided to the radiofrequency power amplifier 2540 based on the values of thetransmit-target power setting parameter 2502A and the transmit-gainsetting parameter 2502B. In one embodiment of the transceiver circuitry2512, the transceiver circuitry 2512 adjusts the transmit-gain settingparameter 2502B in response to changes in bias settings of thetransceiver circuitry 2512. In one embodiment of the transceivercircuitry 2512, the transceiver circuitry 2512 adjusts the transmit-gainsetting parameter 2502B in response to temperature changes of thetransceiver circuitry 2512. In one embodiment of the transceivercircuitry 2512, the transceiver circuitry 2512 adjusts the transmit-gainsetting parameter 2502B in response to changes in the voltage standingwave ratio, VSWR associated with the radio frequency power amplifier2540. In one embodiment of the transceiver circuitry 2512, thetransceiver circuitry 2512 adjusts the transmit-gain setting parameter2502B in response to changes in a transmit frequency of the radiofrequency power amplifier 2540.

In one embodiment of the envelope tracking modulated power supply system2500, the envelope tracking power converter system 2530 generates themodulated power supply voltage, V_(CC), at the modulated power supplyoutput 876 based on the control signal 2508. In general, the modulatedpower supply voltage, V_(CC), is based on the control signal 2508. Inone embodiment of the envelope tracking modulated power supply system2500, the transceiver circuitry 2512 receives the digital I/Q signal2506. In general, the digital I/Q signal 2506 is an I/Q signal. As such,the modulated power supply voltage, V_(CC), is based on the I/Q signaland the modulated radio frequency input signal 2504, which is providedto the radio frequency power amplifier 2540, is also based on the I/Qsignal. In one embodiment of the envelope tracking modulated powersupply system 2500, the transceiver circuitry 2512 adjusts the modulatedpower supply voltage, V_(CC), via the control signal 2508 to compensatefor changes of the transmit-gain setting of the transceiver circuitry2512.

In one embodiment of the envelope tracking modulated power supply system2500, the modulated power supply voltage, V_(CC), is based on the V_(CC)look up table 2586, such that the V_(CC) look up table index signal 2584provides an index signal into the V_(CC) look up table 2586 thatsubstantially emulates the input power P_(IN) of the modulated radiofrequency input signal 2504 provided to the radio frequency poweramplifier 2540. In this regard, the transceiver circuitry 2512 adjuststhe V_(CC) look up table index signal 2584 in response to changes of thetransmit-gain setting of the transceiver circuitry 2512.

In one embodiment of the transceiver circuitry 2512, the transceivercircuitry 2512 adjusts the V_(CC) look up table index signal 2584 inresponse to changes in bias settings of the transceiver circuitry 2512.In one embodiment of the transceiver circuitry 2512, the transceivercircuitry 2512 adjusts the V_(CC) look up table index signal 2584 inresponse to temperature changes of the transceiver circuitry 2512. Inone embodiment of the transceiver circuitry 2512, the transceivercircuitry 2512 adjusts the V_(CC) look up table index signal 2584 inresponse to changes in the voltage standing wave ratio, VSWR associatedwith the radio frequency power amplifier 2540. In one embodiment of thetransceiver circuitry 2512, the transceiver circuitry 2512 adjusts theV_(CC) look up table index signal 2584 in response to changes in atransmit frequency of the radio frequency power amplifier 2540.

In one embodiment of the transceiver circuitry 2512, the transceivercircuitry 2512 further includes the pseudo-envelope follower gaincontrol circuit 2574, which provides the gain control signal 2574A basedon the transmit-gain setting of the transceiver circuitry 2512, suchthat the V_(CC) look up table index signal 2584 is based on the gaincontrol signal 2574A. In one embodiment of the pseudo-envelope followergain control circuit 2574, the gain control signal 2574A is based on adifference between a calibration transmit-gain setting of thetransceiver circuitry 2512 and the transmit-gain setting of thetransceiver circuitry 2512.

In one embodiment of the pseudo-envelope follower gain control circuit2574, the gain control signal 2574A is based on a difference between acalibration temperature of the transceiver circuitry 2512 and a measuredtemperature of the transceiver circuitry 2512. In one embodiment of thepseudo-envelope follower gain control circuit 2574, the gain controlsignal 2574A is based on a difference between a calibration transmitfrequency of the radio frequency power amplifier 2540 and the transmitfrequency of the radio frequency power amplifier 2540. In one embodimentof the pseudo-envelope follower gain control circuit 2574, the gaincontrol signal 2574A is based on the difference between the calibrationtransmit-gain setting of the transceiver circuitry 2512 and thetransmit-gain setting of the transceiver circuitry 2512, the differencebetween the calibration temperature of the transceiver circuitry 2512and the measured temperature of the transceiver circuitry 2512, thedifference between the calibration transmit frequency of the radiofrequency power amplifier 2540 and the transmit frequency of the radiofrequency power amplifier 2540, or any combination thereof.

As depicted in the FIG. 7B, the measured data for the iso-gain contoursmay have a saw tooth characteristic. As depicted in FIG. 7A, the sawtooth effect may be due in part to various corruption sources thataffect the accuracy of calibration including feedback noise sources thatimpact the measured amplified radio frequency output signal 2550,carrier leakage from the oscillator 2558, quadrature mismatch betweenthe in-phase component feedback mixer 2554 and the quadrature componentfeedback mixer 2556, and/or some combination thereof. To correct for thesaw tooth effect, as depicted in FIG. 7B and described in 7C, theinitial measured data may be post-processed either by the controller 50,the baseband controller 2511, or factor test equipment to remove orminimize the impact of the corruption sources. For example, FIG. 7Cdepicts a method 2900 for reducing a saw tooth effect.

At the factor, an initial calibration of each iso-gain contour of anumber of iso-gain contours to be stored in the V_(CC) lookup table 2586may be initiated. (Step 2902.)

For each iso-gain contour of the number of iso-gain contours, the fastcalibration iso-gain and delay alignment circuitry 2524 may store acalibrated transmit-gain setting of the transceiver circuitry 2512, acalibrated measured transceiver temperature, and a calibratedtransmitter frequency. (Step 2904.)

Thereafter, one of the controller 50, the baseband controller 2511, fastcalibration iso-gain and delay alignment circuitry 2524, and/or factortest equipment may post process the initial iso-gain contour data foreach of the iso-gain contours to remove or substantially reduce a sawtooth effect substantially related to various corruption sources in thetransceiver power control loop 2502. As an example, a fit-postprocessing algorithm may be applied to the initial iso-gain contour datato remove the overall saw tooth effect and recalculate discrete pointsfor the V_(CC) look up table 2586 based on an interpolation process. Asan example, the controller 50 or a factory calibration computer mayexecute the fit-processing and then recalculate the new discrete pointto define the iso-gain curves. For example, in some embodiments, N datapoint may be used to generate an M^(th) order polynomial that provides apolyfit with respect to the initial iso-gain contour data. Other advancefiltering techniques may also be applied to remove the saw tooth effectand provide clean post processed calibrated iso-gain contour data. (Step2906).

For each iso-gain contour, one of the controller 50, the basebandcontroller 2511, fast calibration iso-gain and delay alignment circuitry2524, and/or factor test equipment may the store the post-processed(smoothed/interpolated/filtered) initial iso-gain contour calibrationdata as calibrated iso-gain contour calibration data in the V_(CC) lookup table 2586. (Step 2908).

In response to receipt of the in-phase feedback component 2562, thein-phase component analog-to-digital converter 2566 provides thedigitized in-phase feedback component I_(MEASURED) 2570 to the fastcalibration iso-gain and delay alignment circuitry 2524 and thetransceiver power control loop circuitry 2526. Similarly, the quadraturecomponent feedback mixer 2556 provides a quadrature feedback component2564 to the quadrature component analog-to-digital converter 2568. Inresponse, the quadrature component analog-to-digital converter 2568provides the digitized quadrature feedback component Q_(MEASURED) 2572to both the fast calibration iso-gain and delay alignment circuitry 2524and the transceiver power control loop circuitry 2526.

The transceiver power control loop circuitry 2526 may determine ameasured output power, P_(OUT) _(—) _(MEASURED), generated by the radiofrequency power amplifier 2540 in response to the modulated radiofrequency input signal 2504 having an input power P_(IN) and the controlsignal 2508. For example, the transceiver power control loop circuitry2526 may calculate the measured output power, P_(OUT) _(—) _(MEASURED),as a function of the digitized in-phase feedback component I_(MEASURED)2570 and the digitized quadrature feedback component Q_(MEASURED) 2572.Based on the measured output power, P_(OUT) _(—) _(MEASURED), and thetransmit-target power setting parameter 2502A, the transceiver powercontrol loop circuitry 2526 may generate a transmit-gain setting signal2526A to minimize a difference between the measured output power,P_(OUT) _(—) _(MEASURED), and the transmit-target power settingparameter 2502A. The transceiver power control loop circuitry 2526 mayalso provide the transmit-gain setting signal 2526A to a basebandcontroller 2511. In addition, in some embodiments, transceiver powercontrol loop circuitry 2526 may provide the measured output power,P_(OUT) _(—) _(MEASURED), to the controller 50 via the control bus 44.In this case, the baseband controller 2511 may provide thetransmit-target power setting parameter 2502A to the transceiver powercontrol loop circuitry 2526. In some embodiments, the pseudo-envelopefollower gain control circuit 2574 may receive an indication of thetransmit-gain setting parameter 2502B of the transceiver from theconfiguration-feedback circuitry 2520 via the transmit-gain settingsignal 2526A.

FIGS. 8A-D depict a method 2700 for compensating a control signalgenerated as part of an envelope tracking system for changes in thedrive level or input power P_(IN) of a modulated radio frequency inputsignal 2504. The digital baseband circuit 2510 may provide the modulatedradio frequency input signal 2504 to the radio frequency power amplifier2540 to generate the amplified radio frequency output signal 2548. Theradio frequency power amplifier 2540 may provide the amplified radiofrequency output signal 2548 to the antenna 2544 for transmission. Aspreviously discussed, the transceiver power control loop 2502 may adjustthe drive level or input power P_(IN) of the modulated radio frequencyinput signal 2504 in response a determination that the output powerP_(OUT) of the amplified radio frequency output signal 2548, measured atthe antenna 2544, outside a threshold limit of a target-transmit outputpower parameter. For example, the configuration-feedback circuitry 2520may determine whether an output power of a radio frequency poweramplifier is outside of a threshold limit of a target-transmit outputpower parameter. Based on a determination that the output power of theradio frequency power amplifier is outside of the threshold limit of thetarget-transmit output power parameter, a transceiver circuit configuredto generate a modulated radio frequency input signal based on adifference between the output power of the radio frequency poweramplifier and the target-transmit output power parameter, and generate acontrol signal to regulate generation of a modulated supply voltage tomaintain a linear gain characteristic of the radio frequency poweramplifier through a new dynamic range of the modulated radio frequencyinput signal.

The envelope tracking control system 2530 may be configured tocompensate for a change in the input power P_(IN) or drive level of themodulated radio frequency input signal provided to the radio frequencypower amplifier 2540. (Step 2702). As an example, the pseudo-envelopefollower gain control circuit 2574 may receive an indication of thetransmit-gain setting parameter of the transceiver circuit and theconfiguration-feedback circuitry 2520 via the transmit-gain settingsignal 2526A. In addition the pseudo-envelope follower gain controlcircuit 2574 may receive an indication of the transmit-target powersetting parameter 2502A from the configuration-feedback circuitry 2520via the transmit-target power setting signal 2526B. (Step 2704).

In response to receipt of the indication of the transmit-gain settingparameter, the pseudo-envelope follower gain control circuit 2574 mayobtain or receive a value of the calibration transmit-gain settingparameter used during calibration of the iso-gain contours based on thetransmit-target power setting parameter. (Step 2706.)

The pseudo-envelope follower gain control circuit 2574 may determine atransmit gain setting delta as a function of the indication of thetransmit-gain setting of the transceiver and the calibrationtransmit-gain setting parameter. The transmit gain setting delta may becalculated by subtracting the calibration transmit-gain settingparameter from the transmit-gain setting parameter. (Step 2708).

The pseudo-envelope follower gain control circuit 2574 may determine acompensation value for the pseudo-envelope following gain control signalas a function of the temperature of the transceiver circuitry 2512, thetransceiver frequency, and/or a combination thereof (Step 2712.)

The pseudo-envelope follower gain control circuit 2574 may determine ameasured temperature for the transceiver circuitry 2512. To determine acompensation value for the pseudo-envelope following gain control signal2574A, the pseudo-envelope follower gain control circuit 2574 mayreceive a measured temperature for the transceiver circuitry 2512 fromthe temperature sensor 2516 via the temperature signal 2516A. (Step2714.)

The pseudo-envelope follower gain control circuit 2574 may determine themeasured calibration temperature of the transceiver circuitry 2512during the calibration of the iso-gain contours during calibration ofthe iso-gain contours stored in the V_(CC) look up table 2586. As anexample, in some embodiments, the baseband controller 2511 may obtain orreceive the measured calibration temperature of the transceivercircuitry 2512 from the controller 50. Alternatively, the measuredcalibration temperature of the transceiver circuitry 2512 may be storedin a memory or register accessible by the baseband controller 2511. Insome embodiments, the controller 50 may provide the measured calibrationtemperature of the transceiver circuitry 2512 to the pseudo-envelopefollower gain control circuit 2574. In still other embodiments, thepseudo-envelope follower gain control circuit 2574 may request themeasured calibration temperature of the transceiver circuitry 2512 fromthe baseband controller 2511 or the controller 50. In still otherembodiments, the measured calibration temperature of the transceivercircuitry 2512 may be provided to the pseudo-envelope follower gaincontrol circuit 2574 prior to commencement of a data transmission. (Step2716.)

The pseudo-envelope follower gain control circuit 2574 may subtract themeasured calibration temperature from the measured transceivertemperature to determine a transceiver temperature delta. (Step 2718.)

Thereafter, the pseudo-envelope follower gain control circuit 2574 maydetermine a temperature induced transceiver circuitry gain change basedon the transceiver temperature delta and a known temperature dependenttransceiver gain change characteristic/function for the transceivercircuitry 2512. (Step 2720.)

In some embodiments of the envelope tracking control system 2503, thepseudo-envelope follower gain control circuit 2574 may furthercompensate the transceiver frequency being used to generate themodulated radio frequency input signal 2504. In other embodiments, theenvelope tracking control system 2503 may determine whether tocompensate for the transceiver frequency. For example, the envelopetracking control system 2503 may determine whether the presenttransceiver frequency is substantially near a calibration transceiverfrequency used during calibration of the iso-gain contours stored in theV_(CC) look up table 2586. Based on a determination that the presenttransceiver frequency is substantially near the calibration transceiverfrequency, the pseudo-envelope follower gain control circuit 2574 maydetermine to dispense with determining a compensation factor for adifference between the present transceiver frequency and the calibrationtransceiver frequency. In other embodiments, the pseudo-envelopefollower gain control circuit 2574 may always generate a compensationfactor for a difference between the present transceiver frequency andthe calibration transceiver frequency. (Step 2722.)

Based on a determination to not generate a compensation factor for adifference between the present transceiver frequency and the calibrationtransceiver frequency, the pseudo-envelope follower gain control circuit2574 may subtract the temperature induced transceiver circuitry gainchange from the transceiver gain delta to generate the pseudo-envelopefollower gain control signal. (Step 2724.)

Otherwise, based on a determination to generate a compensation factorfor a difference between the present transceiver frequency and thecalibration transceiver frequency, the pseudo-envelope follower gaincontrol circuit 2574 may determine compensation value for transceiverfrequency as a function of the transceiver frequency and the calibrationtransceiver frequency used during calibration of the iso-gain contoursstored in the stored in the V_(CC) look up table 2586. (Step 2726.)

The pseudo-envelope follower gain control circuit 2574 may obtain thetransceiver frequency to be used during the data transmission. Forexample, in some embodiments, the pseudo-envelope follower gain controlcircuit 2574 may obtain or receive the transceiver frequency to be usedduring the data transmission from the baseband controller 2511, thecontroller 50, or the digital baseband circuit 2510. For example, thetransceiver frequency may be based on the band of operation of thecommunication device. The controller 50 or the baseband controller 2511may configure the oscillator 2558 to generate a carrier signal 2558Ahaving a transceiver frequency substantially equal to the transceiverfrequency to be used during the data transmission. (Step 2728.)

The pseudo-envelope follower gain control circuit 2574 may obtain thecalibration transceiver frequency used during calibration of theiso-gain contours stored in the V_(CC) look up table 2586. For example,the pseudo-envelope follower gain control circuit 2574 may obtain orreceive the calibration transceiver frequency used during calibration ofthe iso-gain contours from the controller 50, the baseband controller2511, or from a memory local to the digital baseband circuit 2510. Insome embodiments, the pseudo-envelope follower gain control circuit 2574may receive the calibration transceiver frequency used duringcalibration of the iso-gain contours prior to initiation of a datatransmission. (Step 2730.) The pseudo-envelope follower gain controlcircuit 2574 may subtract the calibration transceiver frequency from thetransceiver frequency to be used during transmission to determine atransceiver frequency delta. (Step 2732.)

Thereafter, the pseudo-envelope follower gain control circuit 2574 maydetermine a transceiver frequency induced gain change based on thetransceiver frequency delta and a known transceiver gain changecharacteristic/function for the transceiver circuitry with respect to achange in the transceiver frequency. (Step 2734.) The pseudo-envelopefollower gain control circuit 2574 may subtract the temperature inducedtransceiver circuitry gain change and the transceiver frequency inducedgain change from the transmit-gain setting delta to generate thepseudo-envelope follower gain control signal. (Step 2736.)

The pseudo-envelope follower gain control circuit 2574 may scale theenvelope magnitude as a function of the pseudo gain control signal togenerate an index input signal for the V_(CC) look up table 2586 thatprovides an emulation of the input power P_(IN) of the modulated radiofrequency input signal provided to the radio frequency power amplifier.(Step 2738.)

The pseudo-envelope follower gain control circuit 2574 may generate acontrol signal to substantially operate on a single iso-gain contourrelated to the transmit-target output power setting parameter tominimize distortion in response to an increase in the input power P_(IN)of the modulated radio frequency input signal provided to the radiofrequency power amplifier (Step 2740.). Thereafter, the digital basebandcircuit 2510 may commence the data transmission. (Step 2742.)

Those skilled in the art will recognize improvements and modificationsto the embodiments of the present disclosure. All such improvements andmodifications are considered within the scope of the concepts disclosedherein and the claims that follow.

What is claimed is:
 1. Circuitry comprising: configuration-feedback circuitry configured to regulate an output power from a radio frequency power amplifier based on a difference between a target output power from the radio frequency power amplifier and a measured output power from the radio frequency power amplifier; and transceiver circuitry configured to regulate a modulated power supply voltage, which is used by the radio frequency power amplifier to provide power for amplification, based on the difference between the target output power from the radio frequency power amplifier and the measured output power from the radio frequency power amplifier, wherein: the transceiver circuitry is further configured to provide a modulated radio frequency input signal to the radio frequency power amplifier; the transceiver circuitry comprises a V_(CC) look up table; the modulated power supply voltage is based on the V_(CC) look up table; and a V_(CC) look up table index signal is configured to provide an index signal into the V_(CC) look up table that substantially emulates an input power of the modulated radio frequency input signal.
 2. The circuitry of claim 1 wherein the configuration-feedback circuitry is further configured to regulate the output power from the radio frequency power amplifier, such that the measured output power from the radio frequency power amplifier is about equal to the target output power from the radio frequency power amplifier.
 3. The circuitry of claim 1 wherein: the transceiver circuitry is further configured to provide a modulated radio frequency input signal having an input power to the radio frequency power amplifier; the radio frequency power amplifier is configured to generate an amplified radio frequency output signal based on the input power of the modulated radio frequency input signal and the modulated power supply voltage; the measured output power from the radio frequency power amplifier is representative of the output power from the radio frequency power amplifier; and a measured amplified radio frequency output signal is representative of the measured output power from the radio frequency power amplifier.
 4. The circuitry of claim 1 further comprising the radio frequency power amplifier.
 5. The circuitry of claim 1 wherein the configuration-feedback circuitry is further configured to: receive a measured amplified radio frequency output signal; provide a transmit-target power setting signal, which is based on a transmit-target power setting parameter; and provide a transmit-gain setting signal, which is based on a transmit-gain setting parameter.
 6. The circuitry of claim 5 wherein: the transmit-target power setting parameter is representative of the target output power from the radio frequency power amplifier; and the transmit-gain setting parameter is representative of a transmit-gain setting of the transceiver circuitry.
 7. The circuitry of claim 5 wherein the transmit-gain setting parameter is based on the measured amplified radio frequency output signal and the transmit-target power setting parameter.
 8. The circuitry of claim 5 wherein the transceiver circuitry is further configured to provide and adjust an input power of a modulated radio frequency input signal to the radio frequency power amplifier based on the transmit-target power setting parameter and the transmit-gain setting parameter.
 9. The circuitry of claim 5 wherein the transceiver circuitry is further configured to adjust the transmit-gain setting parameter in response to changes in bias settings of the transceiver circuitry.
 10. The circuitry of claim 5 wherein the transceiver circuitry is further configured to adjust the transmit-gain setting parameter in response to temperature changes of the transceiver circuitry.
 11. The circuitry of claim 5 wherein the transceiver circuitry is further configured to adjust the transmit-gain setting parameter in response to changes in a voltage standing wave ratio associated with the radio frequency power amplifier.
 12. The circuitry of claim 5 wherein the transceiver circuitry is further configured to adjust the transmit-gain setting parameter in response to changes in a transmit frequency of the radio frequency power amplifier.
 13. The circuitry of claim 1 wherein an envelope tracking power converter system is configured to generate the modulated power supply voltage based on a control signal.
 14. The circuitry of claim 13 further comprising the envelope tracking power converter system.
 15. The circuitry of claim 1 wherein: the transceiver circuitry is further configured to receive an I/Q signal and provide a modulated radio frequency input signal to the radio frequency power amplifier; the modulated power supply voltage is based on the I/Q signal; and the modulated radio frequency input signal is based on the I/Q signal.
 16. The circuitry of claim 1 wherein: the modulated power supply voltage is based on a control signal; and the transceiver circuitry is further configured to adjust the modulated power supply voltage via the control signal to compensate for changes of a transmit-gain setting of the transceiver circuitry.
 17. The circuitry of claim 1 wherein the transceiver circuitry is further configured to adjust the V_(CC) look up table index signal in response to changes in bias settings of the transceiver circuitry.
 18. The circuitry of claim 1 wherein the transceiver circuitry is further configured to adjust the V_(CC) look up table index signal in response to temperature changes of the transceiver circuitry.
 19. The circuitry of claim 1 wherein the transceiver circuitry is further configured to adjust the V_(CC) look up table index signal in response to changes in a voltage standing wave ratio associated with the radio frequency power amplifier.
 20. The circuitry of claim 1 wherein the transceiver circuitry is further configured to adjust the V_(CC) look up table index signal in response to changes in a transmit frequency of the radio frequency power amplifier.
 21. The circuitry of claim 1 wherein the transceiver circuitry is further configured to adjust the V_(CC) look up table index signal in response to changes of a transmit-gain setting of the transceiver circuitry.
 22. The circuitry of claim 1 wherein the transceiver circuitry further comprises a pseudo-envelope follower gain control circuit, which is configured to provide a gain control signal based on a transmit-gain setting of the transceiver circuitry, such that the V_(CC) look up table index signal is based on the gain control signal.
 23. The circuitry of claim 22 wherein the gain control signal is further based on a difference between a calibration transmit-gain setting of the transceiver circuitry and the transmit-gain setting of the transceiver circuitry.
 24. The circuitry of claim 22 wherein the gain control signal is further based on a difference between a calibration temperature of the transceiver circuitry and a measured temperature of the transceiver circuitry.
 25. The circuitry of claim 22 wherein the gain control signal is further based on a difference between a calibration transmit frequency of the radio frequency power amplifier and a transmit frequency of the radio frequency power amplifier.
 26. The circuitry of claim 22 wherein the gain control signal is further based on at least one of a difference between a calibration transmit-gain setting of the transceiver circuitry and the transmit-gain setting of the transceiver circuitry, the difference between a calibration temperature of the transceiver circuitry and a measured temperature of the transceiver circuitry, and a difference between a calibration transmit frequency of the radio frequency power amplifier and a transmit frequency of the radio frequency power amplifier.
 27. A method comprising: regulating an output power from a radio frequency power amplifier based on a difference between a target output power from the radio frequency power amplifier and a measured output power from the radio frequency power amplifier; regulating a modulated power supply voltage, which is used by the radio frequency power amplifier to provide power for amplification, based on the difference between the target output power from the radio frequency power amplifier and the measured output power from the radio frequency power amplifier; providing a modulated radio frequency input signal to the radio frequency power amplifier; providing a V_(CC) look up table, wherein the modulated power supply voltage is based on the V_(CC) look up table; and providing an index signal into the V_(CC) look up table that substantially emulates an input power of the modulated radio frequency input signal. 